Control device for direct power converter for reduction of harmonic distortion

ABSTRACT

A duty calculating unit receives a phase, an amplitude, a command value of a voltage across a capacitor, and a command value of a DC voltage, and calculates an original discharge duty and an original rectification duty. A duty correcting unit corrects the original discharge duty and the original rectification duty to obtain a discharge duty and a rectification duty. The duty calculating unit and the duty correcting unit can collectively be regarded as a duty generating unit that generates the discharge duty and the rectification duty.

TECHNICAL FIELD

The present invention relates to a technique for controlling a directpower converter.

BACKGROUND ART

Power obtained from a single-phase AC power supply contains a componentthat ripples at a frequency twice a power supply frequency. Thus, alarge-capacitance energy accumulation element is required to obtain aconstant DC voltage by using a rectifying circuit.

In response to such a requirement, there has been proposed a techniquefor connecting a capacitor constituting an active buffer to a DC linkthrough a switching element, thereby constituting a voltage source. Sucha configuration forms a high-frequency link together with a power supplyvoltage and achieves high-efficiency characteristics with a sinusoidalinput current in a direct power conversion circuit (for example,Japanese Patent No. 5804167, Japanese Patent No. 5874800 and Yamashita,Sakakibara, “A control method of a single-phase-to-three-phase powerconverter with an active buffer for increasing voltage transfer ratio”,the Institute of Electrical Engineers of Japan Industry ApplicationsSociety Conference 2016, 1-54, pp. I-181 to I-186, 2016).

In this technique, the waveform of a current flowing through the activebuffer (hereinafter also referred to as a “buffer current”) is basicallysinusoidal. The control of such a buffer current is described in, forexample, Japanese Patent No. 5874800 and Yamashita, Sakakibara, “Acontrol method of a single-phase-to-three-phase power converter with anactive buffer for increasing voltage transfer ratio”, the Institute ofElectrical Engineers of Japan Industry Applications Society Conference2016, 1-54, pp. I-181 to I-186, 2016. A circuit that supplies a buffercurrent performs proportional-integral (PI) control on a deviationbetween an actual value of a capacitor voltage and a command value ofthe capacitor voltage. The amplitude of the buffer current is determinedon the basis of the waveform of an input current and the buffer currentis realized by a switching operation such as a critical mode and acontinuous mode.

Saito, “A Single to Three Phase Matrix Converter for Vector ControlledInduction Motor”, the Institute of Electrical Engineers of JapanIndustry Applications Society Conference 2007, 1-O4-5, pp. I-103 toI-108, 2007 proposes a scheme based on a matrix converter as asingle-phase to three-phase conversion circuit including a buffercircuit. The scheme requires a large circuit scale because the scheme isbased on three-phase to three-phase conversion, but is characterized inthat the responsibility for taking measures against noise for a PWMrectifier is significantly decreased, as described in Sakakibara andfour others, “Application of an Indirect Matrix Converter for AirConditioners”, The transactions of the Institute of Electrical Engineersof Japan D, Vol. 136, No. 7, pp. 471-478, 2016.

The above-described scheme using an active buffer requires a simpleconfiguration, but the charge circuit involves a switching operation. Acomponent needs to be added to take measures against noise resultingfrom the switching operation, which may become a factor of degrading thecharacteristic of a simple configuration.

Suga, Kimata, Uchida, “A Simple Switching Method for A Improved PowerFactor Type Single Phase Converter”, The transactions of the Instituteof Electrical Engineers of Japan D, Vol. 116, No. 4, pp. 420-426, 1996and Uesgi and four others, “Single-Phase Twice voltage PFC Converter forair conditioner”, The transactions of the Institute of ElectricalEngineers of Japan D, Vol. 119, No. 5, pp. 592-598, 1999 proposeswitching for simply controlling a charging current (hereinafter alsoreferred to as “simple switching” according to Suga, Kimata, Uchida, “ASimple Switching Method for A Improved Power Factor Type Single PhaseConverter”, The transactions of the Institute of Electrical Engineers ofJapan D, Vol. 116, No. 4, pp. 420-426, 1996, called “partial switching”in Uesgi and four others, “Single-Phase Twice voltage PFC Converter forair conditioner”, The transactions of the Institute of ElectricalEngineers of Japan D, Vol. 119, No. 5, pp. 592-598, 1999). In simpleswitching, the number of switching operations is smaller than that in ascheme using a conventional active buffer, and thus the occurrence ofnoise is reduced.

Japanese Patent No. 5794273, Japanese Patent Application Laid-Open No.2016-103961 and Japanese Patent No. 5772915 describe known techniqueswhich will be described in an embodiment.

SUMMARY OF INVENTION Problems to be Solved by the Invention

However, the waveform of a charging current obtained through simpleswitching is greatly distorted from a sinusoidal waveform. In the priorart introduced above, the waveform of a charging current is basicallysinusoidal. If a charging current obtained through simple switching isused as is, the waveform of an input current is greatly distorted.

Accordingly, an object of the present invention is to provide atechnique for maintaining the function of a power buffer and reducingdistortion of the waveform of an input current from a sinusoidalwaveform even when the waveform of a buffer current is distorted from asinusoidal waveform.

Means to Solve the Problems

The present invention is a control device (10) for a direct powerconverter, which controls the direct power converter. The direct powerconverter includes a DC link (7) including a first DC power supply line(LH; LH1, LH2) and a second DC power supply line; a converter (3) thatreceives a single-phase AC voltage (Vin) and outputs a ripple power(Pin) to the DC link; a power buffer circuit (4) that supplies andreceives a power (Pc, Pl) to and from the DC link; and an inverter (5)that converts a DC voltage between the first DC power supply line andthe second DC power supply line to an AC voltage.

The converter (3) applies a rectified voltage (Vrec) obtained byfull-wave rectifying the single-phase AC voltage (Vin) to the DC link(7) such that the first DC power supply line (LH; LH1, LH2) is higher inpotential than the second DC power supply line (LL).

According to a first aspect, the control device includes a dutygenerating unit (1021, 1023) that generates a rectification duty (drec′,drec″) which is a duty at which a first current (irec1) flows from theconverter to the DC link and a discharge duty (dc′, dc″) which is a dutyat which a second current (ic) flows from the power buffer circuit tothe DC link; and an inverter controller (101) that outputs an invertercontrol signal (SSup, SSvp, SSwp, SSun, SSvn, SSwn) that controls anoperation of the inverter on the basis of the rectification duty, thedischarge duty, and a command value (Vu*, Vv*, Vw*) of a voltage outputfrom the inverter.

The rectification duty is set on the basis of a product of a first ratio(Vdc/Vm) of a predetermined voltage (Vdc) to an amplitude (Vm) of the ACvoltage and an absolute value (|sin(ωt)|) of a sinusoidal value of aphase (ωt) of the AC voltage, and a second ratio (il/Idc, il′/Idc) of athird current (il, il′) to a DC current (Idc) flowing through theinverter. The third current is a fourth current (il) input to the powerbuffer circuit from the DC link or an amount of reduction (il′) of aharmonic component of the fourth current.

A second aspect and a third aspect of the control device for the directpower converter according to the present invention are related to thefirst aspect, and in the second aspect, the converter (3) applies therectified voltage (Vrec) obtained by full-wave rectifying thesingle-phase AC voltage (Vin) to the DC link (7) such that the first DCpower supply line (LH; LH1, LH2) is higher in potential than the secondDC power supply line (LL). The power buffer circuit (4) includes adischarge circuit (4 a) including a capacitor (C4) and a first switch(Sc, D42) that is connected in series to the capacitor between the firstDC power supply line and the second DC power supply line and is closerto the first DC power supply line than the capacitor is, and a chargecircuit (4 b) that charges the capacitor.

According to the second aspect, the control device outputs a dischargeswitch signal (SSc) that brings the first switch into conduction on thebasis of the discharge duty (dc′), and the discharge duty is set on thebasis of a product of a third ratio (Vdc/Vc) of the predeterminedvoltage (Vdc) to a voltage (Vc, Vc*) across the capacitor and a cosinevalue (cos(2ωt)) of a value twice the phase (ωt) of the AC voltage, anda product of a fourth ratio (Vm/Vc) of the amplitude (Vm) to the voltageacross the capacitor, the second ratio (il/Idc), and the absolute value(|sin(ωt)|) of the sinusoidal value. The third current is the fourthcurrent (il).

According to the third aspect, the control device outputs a dischargeswitch signal (SSc) that brings the first switch into conduction on thebasis of the discharge duty (dc″), and the discharge duty is set on thebasis of a product of a third ratio (Vdc/Vc) of the predeterminedvoltage (Vdc) to a voltage (Vc, Vc*) across the capacitor and a square(cos²(ωt)) of a cosine value of the phase (ωt) of the AC voltage, and aproduct of a fourth ratio (Vm/Vc) of the amplitude (Vm) to the voltageacross the capacitor, the second ratio (il′/Idc), and the absolute value(|sin(ωt)|) of the sinusoidal value. The third current is the amount ofreduction (il′) of the harmonic component of the fourth current (il).

For example, the control device (10) further includes anamount-of-correction generating unit (1025) that obtains the amount ofreduction (il′) on the basis of a harmonic component of an input current(Iin) input to the converter (3). Alternatively, the control device (10)further includes an amount-of-correction generating unit (1026) thatobtains the amount of reduction (il′) on the basis of the harmoniccomponent of the fourth current (il).

For example, the duty generating unit includes a duty calculating unit(1021) that obtains an original rectification duty (drec), which is aproduct of the first ratio (Vdc/Vm) and the absolute value (|sin(ωt)|)of the sinusoidal value, and an original discharge duty (dc), which is aproduct of the third ratio (Vdc/Vc) and a square (cos²(ωt)) of a cosinevalue of the phase (ωt) of the AC voltage, and a duty correcting unit(1023) that performs correction based on the second ratio (il/Idc) toobtain the rectification duty (drec′, drec″) from the originalrectification duty and the discharge duty (dc′, dc″) from the originaldischarge duty.

A fourth aspect of the control device for the direct power converteraccording to the present invention is related to the second aspect orthe third aspect, and in the fourth aspect, the charge circuit (4 b)includes a diode (D40) including a cathode connected to the capacitor(C4) and an anode, a reactor (L4) connected between the first DC powersupply line (LH; LH1, LH2) and the anode, and a second switch (Sl, D41)connected between the second DC power supply line (LL) and the anode.According to the third aspect, the control device further includes aswitch control signal generating unit (1031) that generates a controlsignal (SSl) that causes the second switch to be turned on once andturned off once in one period of the rectified voltage (Vrec).

Effects of the Invention

According to the first aspect of the control device for the direct powerconverter according to the present invention, distortion of the waveformof an input current from a sinusoidal waveform is reduced regardless ofthe waveform of the third current.

According to the second aspect or the third aspect of the control devicefor the direct power converter according to the present invention, poweroutput from the inverter is smoothed regardless of the waveform of thethird current.

According to the fourth aspect of the control device for the directpower converter according to the present invention, noise generated bythe second switch is reduced.

The objects, features, aspects, and advantages of the present inventionwill become more apparent from the detailed description given below andthe attached drawings.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram illustrating the configuration of a directpower converter to which a control technique according to the presentinvention can be applied;

FIG. 2 is a block diagram exemplifying the configuration of a controldevice;

FIG. 3 is a block diagram schematically illustrating the input andoutput of power to and from the direct power converter;

FIG. 4 is an equivalent circuit diagram of the direct power converter;

FIG. 5 is a block diagram illustrating an example of a configuration forperforming control to make ripple of output power zero;

FIG. 6 includes graphs showing an operation of the direct powerconverter;

FIG. 7 is a graph showing the waveform of a buffer current when simpleswitching is adopted;

FIG. 8 includes graphs showing an operation of the direct powerconverter;

FIG. 9 includes graphs showing an operation of the direct powerconverter;

FIG. 10 includes graphs showing an operation of the direct powerconverter;

FIG. 11 is a graph showing a relationship between inductance and powerfactor;

FIG. 12 includes graphs showing an operation of the direct powerconverter;

FIG. 13 includes graphs showing an operation of the direct powerconverter;

FIG. 14 includes graphs showing an operation of the direct powerconverter;

FIG. 15 is a graph showing the waveform of a reactor current in ahalf-period of a single-phase AC voltage;

FIG. 16 is a graph showing the waveform of an input current;

FIG. 17 includes graphs showing an operation of the direct powerconverter;

FIG. 18 includes graphs showing an operation of the direct powerconverter;

FIG. 19 includes graphs showing an operation of the direct powerconverter;

FIG. 20 is a graph showing the waveform of a reactor current in ahalf-period of a single-phase AC voltage;

FIG. 21 is a graph showing the waveform of an input current;

FIG. 22 includes graphs showing an operation of the direct powerconverter;

FIG. 23 includes graphs showing an operation of the direct powerconverter;

FIG. 24 is a block diagram exemplifying a first configuration of adischarge controller and its vicinity;

FIG. 25 is a block diagram exemplifying the configuration of anamount-of-correction generating unit;

FIG. 26 is a block diagram exemplifying a second configuration of thedischarge controller and its vicinity;

FIG. 27 is a block diagram exemplifying the configuration of anamount-of-correction generating unit;

FIG. 28 is a circuit diagram illustrating a first modification;

FIG. 29 is a circuit diagram illustrating the configuration on asingle-phase AC power supply side respect to an inverter in a case wherea filter is disposed on a converter and the inverter; and

FIG. 30 is a circuit diagram illustrating the configuration on thesingle-phase AC power supply side respect to the inverter in a casewhere the filter is disposed between the converter and the inverter.

DESCRIPTION OF EMBODIMENTS A. Configurations of Power Converter andControl Device Therefor

FIG. 1 is a block diagram illustrating the configuration of a directpower converter to which a control technique according to the presentinvention can be applied. The direct power converter includes aconverter 3, a power buffer circuit 4, an inverter 5, and a DC link 7.The power buffer circuit 4 functions as the above-described activebuffer.

The converter 3 is connected to a single-phase AC power supply 1, forexample, through a filter 2. The filter 2 includes a reactor L2 and acapacitor C2. The reactor L2 is provided between the converter 3 and oneof two output terminals of the single-phase AC power supply 1. Thecapacitor C2 is provided between the two output terminals of thesingle-phase AC power supply 1. The filter 2 removes, from a current, ahigh-frequency component mainly derived from a switching operation ofthe inverter 5. The filter 2 may be omitted or may be provided betweenthe converter 3 and the power buffer circuit 4. The position of thefilter 2 will be described below in a second modification. For the sakeof simplicity, the function of the filter 2 will be disregarded in thefollowing description.

The DC link 7 includes DC power supply lines LH and LL.

The converter 3 adopts, for example, a diode bridge, and includes diodesD31 to D34. The diodes D31 to D34 constitute a bridge circuit thatsingle-phase full-wave rectifies a single-phase AC voltage Vin, which isan input voltage input from the single-phase AC power supply 1, toconvert the single-phase AC voltage Vin to a rectified voltage Vrec(=|Vin|; Vin=Vm·sin(ωt)), and outputs the rectified voltage Vrec betweenthe DC power supply lines LH and LL. A potential applied to the DC powersupply line LH is higher than a potential applied to the DC power supplyline LL. An input current Iin flows from the single-phase AC powersupply 1 into the converter 3. The converter 3 outputs a current irec(=|Iin|; Iin=Im·sin(ωt)).

The power buffer circuit 4 includes a discharge circuit 4 a, a chargecircuit 4 b, and a current blocking circuit 4 c, and supplies power toand receives power from the DC link 7. The discharge circuit 4 aincludes a capacitor C4 as a buffer capacitor. The charge circuit 4 bboosts the rectified voltage Vrec to charge the capacitor C4. Thecurrent blocking circuit 4 c blocks a current flowing from the dischargecircuit 4 a toward the charge circuit 4 b.

The discharge circuit 4 a further includes a diode D42 and a transistor(here, an insulated gate bipolar transistor: hereinafter abbreviated as“IGBT”) Sc connected in antiparallel to the diode D42. The transistor Scis connected in series to the capacitor C4 between the DC power supplylines LH and LL and is closer to the DC power supply line LH than thecapacitor C4 is.

Here, antiparallel connection means parallel connection in which forwarddirections are opposite to each other. Specifically, the forwarddirection of the transistor Sc is a direction from the DC power supplyline LL toward the DC power supply line LH, and the forward direction ofthe diode D42 is a direction from the DC power supply line LH toward theDC Power supply line LL. The transistor Sc and the diode D42 cancollectively be regarded as a single switch element (switch Sc).Conduction of the switch Sc causes the capacitor C4 to discharge andsupply power to the DC link 7.

The charge circuit 4 b includes, for example, a diode D40, a reactor L4,and a transistor (here, an IGBT) Sl. The diode D40 includes a cathodeand an anode. The cathode is connected between the switch Sc and thecapacitor C4. This configuration is known as a so-called boost chopper.

The reactor L4 is connected between the DC power supply line LH and theanode of the diode D40. The transistor Sl is connected between the DCpower supply line LL and the anode of the diode D40. The transistor Slis connected in antiparallel to a diode D41. The transistor Sl and thediode D41 can collectively be regarded as a single switch element(switch Sl). Specifically, the forward direction of the transistor Sl isa direction from the DC power supply line LH toward the DC power supplyline LL, and the forward direction of the diode D41 is a direction fromthe DC power supply line LL toward the DC power supply line LH.

The capacitor C4 is charged by the charge circuit 4 b to generate avoltage Vc across the capacitor C4, the voltage Vc being higher than therectified voltage Vrec. Specifically, a current is caused to flow fromthe DC power supply line LH to the DC power supply line LL via theswitch Sl to accumulate energy in the reactor L4. Subsequently, theswitch Sl is turned off, so that the energy is accumulated in thecapacitor C4 via the diode D40.

Since the voltage Vc across the capacitor C4 is higher than therectified voltage Vrec, no current basically flows through the diodeD42. Thus, whether or not the switch Sc is conducting depends solely onwhether or not the transistor Sc is conducting. Here, the diode D42ensures a reverse breakdown voltage when the voltage Vc across thecapacitor C4 is lower than the rectified voltage Vrec, and functions tocause a current flowing back from an inductive load 6 to the DC link 7to perform reverse conduction when the inverter 5 abnormally stops.

Furthermore, since the DC power supply line LH is higher in potentialthan the DC power supply line LL, no current basically flows through thediode D41. Thus, whether or not the switch Sl is conducting dependssolely on whether or not the transistor Sl is conducting. Here, thediode D41 is exemplified as a diode for bringing a reverse breakdownvoltage and reverse conduction and as a diode built in the transistor Slwhich is an IGBT, but the diode D41 does not play a part in a circuitoperation.

The current blocking circuit 4 c is provided on the DC power supply lineLH between the charge circuit 4 b and the discharge circuit 4 a. Forexample, a diode D43 serves as the current blocking circuit 4 c. Theanode of the diode D43 is connected to the reactor L4 on the oppositeside of the switch Sl (i.e., on the converter 3 side). The cathode ofthe diode D43 is connected to the switch Sc on the opposite side of thecapacitor C4 (i.e., on the inverter 5 side). The current blockingcircuit 4 c is known from, for example, Japanese Patent No. 5772915.

The inverter 5 converters a DC voltage between the DC power supply linesLH and LL into an AC voltage and outputs the AC voltage to outputterminals Pu, Pv, and Pw. The inverter 5 includes six switching elementsSup, Svp, Swp, Sun, Svn, and Swn. The switching elements Sup, Svp, andSwp are connected between the output terminals Pu, Pv, and Pw,respectively, and the DC power supply line LH. The switching elementsSun, Svn, and Swn are connected between the output terminals Pu, Pv, andPw, respectively, and the DC power supply line LL. The inverter 5 is aso-called voltage source inverter and includes six diodes Dup, Dvp, Dwp,Dun, Dvn, and Dwn.

Each of the diodes Dup, Dvp, Dwp, Dun, Dvn, and Dwn is arranged suchthat the cathode thereof is directed toward the DC power supply line LHand the anode thereof is directed toward the DC power supply line LL.The diode Dup is connected in parallel to the switching element Supbetween the output terminal Pu and the DC power supply line LH.Likewise, the diodes Dvp, Dwp, Dun, Dvn, and Dwn are connected inparallel to the switching elements Svp, Swp, Sun, Svn, and Swn,respectively. The output terminals Pu, Pv, and Pw output load currentsiu, iv, and iw, respectively. These load currents constitute athree-phase AC current. For example, IGBTs are adopted as the switchingelements Sup, Svp, Swp, Sun, Svn, and Swn.

The inductive load 6 is, for example, a rotary machine, and isillustrated as an equivalent circuit representing an inductive load.Specifically, a reactor Lu and a resistor Ru are connected in series toeach other, and one end of this series structure is connected to theoutput terminal Pu. Similar series structures are obtained for reactorsLv and Lw and resistors Rv and Rw. The other ends of these seriesstructures are connected to each other.

When a control system is exemplified in which the inductive load 6serves as a synchronous machine, a velocity detector 9 detects the loadcurrents iu, iv, and iw flowing through the inductive load 6 andprovides a control device 10 for the direct power converter with arotational angular velocity ωm, a q-axis current Iq, and a d-axiscurrent Id (to be precise, information representing them; the sameapplies hereinafter) that are obtained from the load currents iu, iv,and iw.

The control device 10 receives an amplitude Vm of the single-phase ACvoltage Vin, an angular velocity co (or a phase θ=ωt, which is theproduct of the angular velocity ω and a time t), a command value ωm* ofthe rotational angular velocity, a command value Vq* of a q-axis voltage(hereinafter also referred to as a “q-axis voltage command value”), acommand value Vd* of a d-axis voltage (hereinafter also referred to as a“d-axis voltage command value”), and currents Ish and il, as well as therotational angular velocity ωm, the q-axis current Iq, and the d-axiscurrent Id.

Here, the current Ish is an instantaneous value of a current flowingthrough the inverter 5 and is measured in either the DC power supplyline LL or LH by using a known technique. The current il is a reactorcurrent flowing through the reactor L4 and corresponds to theabove-described buffer current. The reactor current il is measured by,for example, a known current protecting device. The configuration forobtaining the currents Ish and il is a known technique and thus theillustration thereof is omitted here.

FIG. 2 is a block diagram exemplifying the configuration of the controldevice 10. The control device 10 includes an inverter controller 101, adischarge controller 102, and a charge controller 103.

The inverter controller 101 outputs inverter control signals SSup, SSvp,SSwp, SSun, SSvn, and SSwn on the basis of a discharge duty dc′, arectification duty drec′, and command values of voltages (hereinafteralso referred to as “voltage command values”) Vu*, Vv*, and Vw* outputfrom the inverter 5, which will be described below. The inverter controlsignals SSup, SSvp, SSwp, SSun, SSvn, and SSwn control the operations ofthe switching elements Sup, Svp, Swp, Sun, Svn, and Swn, respectively.

The inverter controller 101 includes an output voltage commandgenerating unit 1011 that generates the voltage command values Vu*, Vv*,and Vw* on the basis of the phase θ (=cot), the q-axis current Iq, thed-axis current Id, the rotational angular velocity ωm, and its commandvalue ωm*.

The inverter controller 101 further includes an amplitude modulationcommand unit 1012, a multiply-accumulate operation unit 1013, acomparing unit 1014, and a logical operation unit 1015.

The amplitude modulation command unit 1012 controls an operation of themultiply-accumulate operation unit 1013 on the basis of the dischargeduty dc′ and the rectification duty drec′. The multiply-accumulateoperation unit 1013 (only the signs of multipliers are illustrated forthe sake of simplicity) performs multiply-accumulate operation betweenthe voltage command values Vu*, Vv*, and Vw*, and the discharge duty dc′and the rectification duty drec′, thereby generating signal waves M.

The comparing unit 1014 outputs results of comparison in value betweenthe signal waves M and a carrier CA to the logical operation unit 1015.The logical operation unit 1015 performs logical operation on theresults of comparison and outputs the inverter control signals SSup,SSvp, SSwp, SSun, SSvn, and SSwn.

The discharge controller 102 includes a duty calculating unit 1021, acomparator 1022, and a duty correcting unit 1023.

The duty calculating unit 1021 receives the phase θ, the amplitude Vm, acommand value Vc* of the voltage Vc across the capacitor C4, and acommand value Vdc* of a DC voltage Vdc described below, and calculatesan original discharge duty dc and an original rectification duty drec.

The duty correcting unit 1023 corrects the original discharge duty dcand the original rectification duty drec to obtain the discharge dutydc′ and the rectification duty drec′.

Accordingly, the duty calculating unit 1021 and the duty correcting unit1023 can collectively be regarded as a duty generating unit thatgenerates the discharge duty dc′ and the rectification duty drec′.

The generation of the original discharge duty dc, the originalrectification duty drec, the discharge duty dc′, and the rectificationduty drec′ will be described below.

The comparator 1022 compares the discharge duty dc′ with the carrier CAto generate a discharge switch signal SSc for bringing the switch Scinto conduction.

The operations of the inverter controller 101 and the comparator 1022are known techniques (see, for example, Japanese Patent No. 5804167 andJapanese Patent No. 5874800), and thus the details thereof is omittedhere.

The charge controller 103 includes a switch control signal generatingunit 1031 that generates a control signal SSl for controlling ON and OFFof the switch Sl. The switch control signal generating unit 1031 isknown from Suga, Kimata, Uchida, “A Simple Switching Method for AImproved Power Factor Type Single Phase Converter”, The transactions ofthe Institute of Electrical Engineers of Japan D, Vol. 116, No. 4, pp.420-426, 1996 and Uesgi and four others, “Single-Phase Twice voltage PFCConverter for air conditioner”, The transactions of the Institute ofElectrical Engineers of Japan D, Vol. 119, No. 5, pp. 592-598, 1999, andthus the details thereof is omitted here but will be briefly describedbelow. For example, the switch control signal generating unit 1031generates the control signal SSl on the basis of an output power Pout,which is instantaneous power output from the inverter 5, and theamplitude Vm.

B. Outline of Operation of Power Buffer Circuit 4

A converter input power Pin, which is instantaneous power input to theconverter 3, is expressed by the following Equation (1), with anamplitude Im of the input current Iin being introduced and an inputpower factor being 1.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 1} \right\rbrack & \; \\\begin{matrix}{{Pin} = {{Vm} \cdot {Im} \cdot {\sin^{2}\left( {\omega\; t} \right)}}} \\{= {\frac{{Vm} \cdot {Im}}{2} - {\frac{{Vm} \cdot {Im}}{2} \cdot {\cos\left( {2\;\omega\; t} \right)}}}}\end{matrix} & (1)\end{matrix}$

The converter input power Pin has an AC component (−1/2)·Vm·Im·cos(2ωt)represented by the second term of the rightmost side of Equation (1)(hereinafter also referred to as an “AC component Pin{circumflex over( )}”). Thus, hereinafter the converter input power Pin may also bereferred to as a ripple power Pin.

The power converter illustrated in FIG. 1 can be grasped as follows.

The converter 3 receives the single-phase AC voltage Vin and outputs theripple power Pin. The power buffer circuit 4 receives an instantaneouspower Pl (hereinafter also referred to as a “receiving power Pl”) fromthe DC link 7 and outputs an instantaneous power Pc (hereinafter alsoreferred to as a “supply power Pc”) to the DC link 7. The inverter 5receives, from the DC link 7, an inverter input power Pdc (=Pin+Pc−Pl),which is an instantaneous power obtained by subtracting the receivingpower Pl from the sum of the ripple power Pin and the supply power Pc,and outputs the load currents iu, iv, and iw. When the loss of theinverter 5 is disregarded, the inverter input power Pdc is equal to theoutput power Pout.

FIG. 3 is a block diagram schematically illustrating the input andoutput of power to and from the direct power converter illustrated inFIG. 1. An instantaneous power Pbuf subjected to buffering (hereinafteralso referred to as a “buffering power Pbuf”) is equal to a powerdifference (Pc−Pl) obtained by subtracting the receiving power Pl fromthe supply power Pc. An instantaneous power Prec transmitted from theconverter 3 toward the inverter 5 is equal to a power difference(Pin−Pl). Thus, Pdc=Prec+Pc holds.

When the power buffer circuit 4 supplies and receives the powercorresponding to an absolute value |Pin{circumflex over ( )}| of the ACcomponent Pin′ to and from the DC link 7, Pdc=Pin−Pin{circumflex over( )}, and the following Equation (2) holds.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 2} \right\rbrack & \; \\{{Pdc} = \frac{{Vm} \cdot {Im}}{2}} & (2)\end{matrix}$

C. Equivalent Circuit of Direct Power Converter and Various Duties

FIG. 4 illustrates an equivalent circuit of the direct power converterillustrated in FIG. 1. The equivalent circuit is introduced in, forexample, Japanese Patent No. 5804167 and Japanese Patent No. 5874800. Inthe equivalent circuit, a current irec1 is equivalently represented as acurrent irec1 that flows via a switch Srec when the switch Srec isconducting. Likewise, a discharge current ic is equivalently representedas a current ic that flows via the switch Sc when the switch Sc isconducting.

In addition, a current flowing into the inductive load 6 via theinverter 5 when the output terminals Pu, Pv, and Pw are connected incommon to either the DC power supply line LH or LL in the inverter 5 isequivalently represented as a zero-phase current iz that flows via aswitch Sz when the switch Sz is conducting.

FIG. 4 illustrates the reactor L4, the diode D40, and the switch Sl thatconstitute the charge circuit 4 b, and also illustrates the reactorcurrent il flowing through the reactor L4.

In this equivalent circuit, duties drec′, dc′, and dz′ at which theswitches Srec, Sc, and Sz are brought into conduction, respectively, areintroduced. Note that, as is known from the above-mentioned documents,0≤drec′≤1, 0≤dc′≤1, 0≤dz′≤1, and drec′+dc′+dz′=1.

The duty drec′ is a duty for setting a time period over which theconverter 3 is connected to the DC link 7 and a current is allowed toflow through the inverter 5, and is thus the rectification duty drec′described above.

The duty dc′ is a duty at which the capacitor C4 discharges, and is thusthe discharge duty dc′ described above.

The duty dz′ is a duty at which the zero-phase current iz always flowsregardless of an output voltage in the inverter 5, and thus may bereferred to as a zero duty dz′.

A DC current Idc is a current flowing into the inductive load 6 via theinverter 5 and can be calculated from the current Ish in a mannerdescribed below. The currents irec1, ic, and iz are obtained bymultiplying the duties drec′, dc′, and dz′, respectively, by the DCcurrent Idc. Thus, the currents irec1, ic, and iz are average values inthe switching periods of the switches Srec, Sc, and Sz, respectively.The duties drec′, dc′, and dz′ can also be regarded as currentdistribution factors of the DC current Idc to the currents irec1, ic,and iz, respectively.

In the case of adopting a diode bridge as the converter 3, the converter3 is unable to actively perform switching at the rectification dutydrec′. Thus, the inverter 5 and the switch Sc perform switching inaccordance with the zero duty dz′ and the discharge duty dc′,respectively, thereby being able to obtain the current irec1.

In a time period over which the zero-phase current iz flows, theinverter 5 is unable to use a DC voltage in the DC link 7. Thus, the DCvoltage in the DC link 7, used for supplying power to the inverter 5holds significance in power conversion. In other words, an instantaneousDC voltage that is not used by the inverter 5 for power conversion doesnot hold significance. With the DC voltage Vdc holding significance inpower conversion being introduced and Equation (2) being considered, theDC current Idc can be expressed by the following Equation (3). The DCvoltage Vdc can be expressed by the following Equation (4).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 3} \right\rbrack & \; \\{{Idc} = {\frac{Pdc}{Vdc} = \frac{{Vm} \cdot {Im}}{2 \cdot {Vdc}}}} & (3) \\\left\lbrack {{Math}.\mspace{14mu} 4} \right\rbrack & \; \\{{Vdc} = {{{Vrec} \cdot {drec}^{\prime}} + {{Vc}*{\cdot {dc}^{\prime}}} + {0 \cdot {dz}^{\prime}}}} & (4)\end{matrix}$

On the other hand, the DC voltage Vdc can also be grasped as a voltageto be applied to the DC link 7 as an average value of maximum values ofthe voltage that the inverter 5 can output in the period of controllingswitching of the switches Sc and Sl and the inverter 5. This is becausethe inverter 5 is insulated from either the DC power supply line LL orLH during a time period corresponding to the zero duty dz′ although theinverter 5 may contribute to the voltage of the DC link 7 at a ratiorepresented by the zero duty dz′.

In FIG. 4, the DC voltage Vdc is illustrated as a voltage that isgenerated across a current source Idc (that supplies the DC current Idc)representing the inverter 5 and the inductive load 6.

In the present invention, equations using the amplitude Im of the inputcurrent are used, but it is not always necessary to measure theamplitude Im. For example, the inverter input power Pdc can becalculated in the manner described below.

As an example, a description will be given of the case of performingwell-known control of dq axes regarding an operation of a normal ACload. A power equation on the dq axes is typically expressed by Equation(5). Symbols V* and I represent a command value of a voltage applied tothe AC load and a current flowing through the AC load, respectively.These are AC, and thus a dot representing a complex number is attachedon each of the symbols V* and I. Note that the q-axis voltage ideallyfollows its command value, that is, the q-axis voltage command valueVq*, and the d-axis voltage ideally follows its command value, that is,the d-axis voltage command value Vd*.[Math. 5]P+jQ={dot over (V)}*·İ=Vd*·Id+Vq*·Iq+j(Vq*·Id−Vd*·Iq)  (5)

There is no reactive power in the inverter input power Pdc supplied fromthe DC power supply lines LH and LL to the inverter 5, and thus theinverter input power Pdc is expressed by Equation (6), with the secondterm of the rightmost side of Equation (5) being disregarded.[Math. 6]Pdc=Vd*·Id+Vq*·Iq  (6)

Thus, control of making a ripple (AC component) zero of Equation (6)enables control of realizing Equations (3) and (4) to be performed. Anexample of a configuration for performing the foregoing control isillustrated in FIG. 5 as a block diagram. This configuration is providedin, for example, a configuration illustrated as the output voltagecommand generating unit 1011 in FIG. 2.

A brief description will be given of the part corresponding to a knowntechnique in the configuration in FIG. 5. Trigonometric function valuescos β* and −sin β* are calculated from a current phase command value β*,and a q-axis current command value Iq* and a d-axis current commandvalue Id* are generated from the trigonometric function values cos β*and −sin β* and a current command value Ia*. Assuming that the inductiveload 6 is a rotary machine, the q-axis voltage command value Vq* and thed-axis voltage command value Vd* are calculated on the basis of therotational angular velocity ωm of the rotary machine, a field magneticflux ϕa of the rotary machine, a d-axis inductance Iq* and a q-axisinductance Lq of the rotary machine, the q-axis current command valueIq*, the d-axis current command value Id*, the q-axis current Iq, andthe d-axis current Id. On the basis of the q-axis voltage command valueVq* and the d-axis voltage command value Vd*, the voltage command valuesVu*, Vv*, and Vw* for controlling the inverter 5 are generated.

For example, in the configuration illustrated in FIG. 1, the velocitydetector 9 detects the load currents iu, iv, and iw flowing through theinductive load 6 and supplies the control device 10 with the rotationalangular velocity ωm obtained from the load currents, the q-axis currentIq, and the d-axis current Id.

A DC power calculating unit 711 receives the q-axis voltage commandvalue Vq*, the d-axis voltage command value Vd*, the q-axis current Iq,and the d-axis current Id, calculates the inverter input power Pdc onthe basis of Equation (6) given above, and supplies a calculation resultto a ripple extracting unit 712.

The ripple extracting unit 712 extracts and outputs an AC component ofEquation (6). For example, a high-pass filter HPF serves as the rippleextracting unit 712. A PI processing unit 716 performsproportional-integral control on the AC component, and a value obtainedthereby is output to a subtracter 715.

The subtracter 715 performs a process of correcting the current commandvalue Ia* in a normal process by using the output of the PI processingunit 716. Specifically, as a normal process for calculating the currentcommand value Ia*, a subtracter 701 calculates a deviation between therotational angular velocity ωm and the command value ωm* thereof. Thedeviation is subjected to proportional-integral control by a PIprocessing unit 702, and the current command value Ia* is oncecalculated. Subsequently, the subtracter 715 performs a process ofdecreasing the current command value Ia* with the output from the PIprocessing unit 716.

The above-described known technique is applied to the current commandvalue Ia* corrected in this manner by a processing unit 71, so that theq-axis voltage command value Vq* and the d-axis voltage command valueVd* are generated. With this control, control is performed by givingfeedback on the q-axis voltage command value Vq* and the d-axis voltagecommand value Vd*, and the q-axis current Iq and the d-axis current Id,so that the AC component of the inverter input power Pdc converges tozero.

D. Principle of Present Embodiment

As described in Japanese Patent No. 5804167 and Japanese Patent No.5874800 and so forth, the rectification duty drec′ and the dischargeduty dc′ are determined without particularly drawing a distinctionbetween a reception period and a supply period in the presentembodiment. First, the original rectification duty drec and the originaldischarge duty dc are once defined by Equations (7) and (8),respectively. Note that the command value Vdc* of the DC voltage Vdc canbe used as the DC voltage Vdc. That is, the DC voltage Vdc will behereinafter handled as a predetermined voltage.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 7} \right\rbrack & \; \\{{drec} = {\frac{Vdc}{Vm} \cdot {{\sin\left( {\omega\; t} \right)}}}} & (7) \\\left\lbrack {{Math}.\mspace{14mu} 8} \right\rbrack & \; \\\begin{matrix}{{dc} = {\frac{Vdc}{{Vc}*} \cdot {\cos^{2}\left( {\omega\; t} \right)}}} \\{= {\frac{Vdc}{{2 \cdot {Vc}}*} \cdot \left\lbrack {1 + {\cos\left( {2\;\omega\; t} \right)}} \right\rbrack}}\end{matrix} & (8)\end{matrix}$

As long as Vdc≤Vm, 0≤drec≤1 is satisfied. In addition, a settingsatisfying 0≤dc≤1 is possible when the DC voltage Vdc is set to besmaller than or equal to the command value Vc*.

Equation (9) is obtained from Equations (7) and (8). Equation (9)matches a case where drec′=drec, dc′=dc, and dz′=dz in Equation (4).Thus, Equations (4) and (9) express the validity of control using theoriginal rectification duty drec and the original discharge duty dc.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 9} \right\rbrack & \; \\\begin{matrix}{{{{Vrec} \cdot {drec}} + {{Vc}*{\cdot {dc}}} + {0 \cdot {dz}}} = {{{Vm} \cdot {{\sin\left( {\omega\; t} \right)}} \cdot \frac{Vdc}{Vm} \cdot {{\sin\left( {\omega\; t} \right)}}} +}} \\{{Vdc} \cdot {\cos^{2}\left( {\omega\; t} \right)}} \\{= {Vdc}}\end{matrix} & (9)\end{matrix}$

On the basis of Equations (3) and (7), the current irec1 in the case ofperforming control using the original rectification duty drec iscalculated by using the following Equation (10).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 10} \right\rbrack & \; \\\begin{matrix}{{{irec}\; 1} = {{drec} \cdot {Idc}}} \\{= {\frac{Vdc}{Vm} \cdot {{\sin\left( {\omega\; t} \right)}} \cdot \frac{{Vm} \cdot {Im}}{2 \cdot {Vdc}}}} \\{= {\frac{Im}{2} \cdot {{\sin({\omega t})}}}}\end{matrix} & (10)\end{matrix}$

As described in Japanese Patent No. 5804167 and Japanese Patent No.5874800, when the reactor current il has a value il0=irec1 expressed bythe following Equation (11), the input current Iin becomes sinusoidalwith use of the original rectification duty drec and the originaldischarge duty dc.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 11} \right\rbrack & \; \\\begin{matrix}{{il} = {{{il}\; 0} = {{{Im} \cdot {{\sin\left( {\omega\; t} \right)}}} - {{irec}\; 1}}}} \\{= {\frac{Im}{2} \cdot {{\sin({\omega t})}}}} \\{= {{irec}\; 1}}\end{matrix} & (11)\end{matrix}$

FIG. 6 includes graphs showing an operation of the direct powerconverter illustrated in FIG. 1, and illustrates an operation undercontrol using the original rectification duty drec and the originaldischarge duty dc (dz=1−dc−drec).

In FIG. 6, the top graph shows the duties drec, dc, and dz; the secondgraph from the top shows the DC voltage Vdc, the voltages drec·Vrec anddc·Vc (see Equation (4)) that constitute the DC voltage Vdc, and the DCcurrent Idc; the third graph from the top shows the currents irec, ic,il, and irec1; and the bottom graph shows the instantaneous powers Pin,Pdc, Pbuf, Pc, −Pl, and Prec. It is assumed that the voltage Vc acrossthe capacitor C4 accurately follows the command value Vc*.

In FIG. 6, the horizontal axis represents the phase ωt, with “degrees”being used as the unit. The currents Idc, irec, ic, il, and irec1 areconverted by using an amplitude Im of √2. The voltages Vrec·drec andVc·dc are converted by using an amplitude Vm of 1, and Vc=1.14 Vm.Vdc=0.86 Vm, so that the minimum value of dz=1−dc−drec is 0. Theinstantaneous powers Pin, Pout, Pbuf, Pc, −Pl, and Prec are calculatedas the products of the voltages converted in the above-described mannerand currents.

FIG. 6 exemplifies a case where the reactor current il has a value il0,where the waveform of the current irec exhibits an absolute value of asine wave.

However, if the reactor current il that does not have a waveform of theabsolute value of a sine wave is allowed to flow in the control usingthe original rectification duty drec, the input current Iin may bedistorted from a sinusoidal waveform. Thus, the direct power converteris controlled by using the rectification duty drec′ obtained bycorrecting the original rectification duty drec as described below.

The current irec1 is decreased in response to a requirement ofcancelling out an influence on the input current Iin when the reactorcurrent il is larger than the value il0. Thus, the rectification dutydrec′ is decreased to be smaller than the original rectification dutydrec. Thus, the relationship of the following Equation (12) isestablished in view of the continuity of current. When il=il0,drec′=drec.[Math. 12](drec′−drec)·Idc+(il−il0)=0∴drec′=drec−(il−il0)/Idc  (12)

Deformation of Equation (12) produces the following Equation (13). Thesecond term in the square brackets (the symbol “[” and the symbol “]”)of the rightmost side is a correction term for the originalrectification duty drec. The correction term is a value obtained bysubtracting a ratio (il/Idc) from the original rectification duty drec.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 13} \right\rbrack & \; \\\begin{matrix}{{drec}^{\prime} = {\frac{{irec}\; 1}{Idc} - \frac{{il} - {{irec}\; 1}}{Idc}}} \\{= {{drec} + \left\lbrack {{drec} - \left( \frac{il}{Idc} \right)} \right\rbrack}}\end{matrix} & (13)\end{matrix}$

The adoption of the rectification duty drec′ decreases distortion of theinput current Iin from a sinusoidal waveform regardless of the waveformof the reactor current il.

When the reactor current il that does not have a waveform of theabsolute value of a sine wave is allowed to flow in the control usingthe original discharge duty dc, the waveform of the buffering power Pbufbecomes distorted and the output power Pout (=Pdc) has ripple, althoughthe function of the power buffer is maintained. Thus, the direct powerconverter is controlled by using the discharge duty dc′ obtained bycorrecting the original discharge duty dc as will be described below.

The discharge current ic is increased in response to a requirement ofcancelling out an influence on the voltage Vc across the capacitor C4when the reactor current il is larger than the value il0. Thus, thedischarge duty dc′ is increased to be larger than the original dischargeduty dc. Thus, the relationship of the following Equation (14) isestablished in view of the power for charging the capacitor C4. Whenil=il0, dc′=dc.[Math. 14]Vc*·Idc·(dc′−dc)=(il−il0)·Vm·|sin(ωt)   (14)

Regarding a part of the right side of Equation (14), the followingEquation (15) is obtained.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 15} \right\rbrack & \; \\\begin{matrix}{{{Vm} \cdot \frac{{- {il}}\;{0 \cdot {{\sin\left( {\omega\; t} \right)}}}}{{Vc}*{\cdot {Idc}}}} = {{Vm} \cdot \frac{{- {drec}} \cdot {{\sin\left( {\omega\; t} \right)}}}{{Vc}*}}} \\{= {- \frac{{Vdc} \cdot {{\sin\left( {\omega\; t} \right)}} \cdot {{\sin\left( {\omega\; t} \right)}}}{{Vc}*}}} \\{= {{- \frac{Vdc}{2\;{Vc}*}} \cdot \left\lbrack {1 - {\cos\left( {2\;\omega\; t} \right)}} \right\rbrack}}\end{matrix} & (15)\end{matrix}$

Accordingly, the following Equation (16) is obtained from Equations (14)and (15).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 16} \right\rbrack & \; \\{{dc}^{\prime} = {{\frac{Vdc}{{Vc}*} \cdot {\cos\left( {2\;\omega\; t} \right)}} + {\left( \frac{il}{Idc} \right) \cdot \frac{Vm}{{Vc}*} \cdot {{\sin\left( {\omega\; t} \right)}}}}} & (16)\end{matrix}$

Regarding the first term of the right side of Equation (16), thefollowing equation (17) is obtained.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 17} \right\rbrack & \; \\\begin{matrix}{{\frac{Vdc}{{Vc}*} \cdot {\cos\left( {2\;\omega\; t} \right)}} = {\frac{Vdc}{2\;{Vc}*} \cdot \left\lbrack {2 + {2 \cdot {\cos\left( {2\;\omega\; t} \right)}} - 2} \right\rbrack}} \\{= {{2 \cdot {dc}} - \frac{Vdc}{{Vc}*}}}\end{matrix} & (17)\end{matrix}$

The following Equation (18) is obtained from Equations (16) and (17).The second term in the square brackets (the symbol “[” and the symbol“]”) of the right side is a correction term for the original dischargeduty dc. The correction term is a value obtained by adding a first valueto the original discharge duty dc. The first value is a value obtainedby dividing a second value by the command value Vc*. The second value isa value obtained by subtracting the DC voltage Vdc from the product ofthe ratio (il/Idc) and the rectified voltage Vrec.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 18} \right\rbrack & \; \\{{dc}^{\prime} = {{dc} + \left\lbrack {{dc} + \frac{{\left( \frac{il}{Idc} \right) \cdot {Vm} \cdot {{\sin\left( {\omega\; t} \right)}}} - {Vdc}}{{Vc}*}} \right\rbrack}} & (18)\end{matrix}$

The adoption of the discharge duty dc′ smooths the output power Pout andremoves a ripple component therefrom regardless of the waveform of thereactor current il.

The output power Pout is equal to the product Vdc·Idc, which is equal tothe inverter input power Pdc. Accordingly, the following Equation (19)holds with reference to Equations (2), (9), and (11).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 19} \right\rbrack & \; \\\begin{matrix}{{{{Vrec} \cdot {drec}^{\prime}} + {{Vc}*{\cdot {dc}^{\prime}}}} = {{{Vrec} \cdot \left\lbrack {{drec} - {\left( {{il} - {{il}\; 0}} \right)/{Idc}}} \right\rbrack} +}} \\{{{Vdc} \cdot {\cos\left( {2\;\omega\; t} \right)}} + {\left( {{il}/{Idc}} \right) \cdot}} \\{{Vm} \cdot {{\sin\left( {\omega\; t} \right)}}} \\{= {{{Vrec} \cdot {drec}} - {{Vrec} \cdot \left( {{il}/{Idc}} \right)} +}} \\{{{Vrec} \cdot \left( {{il}\;{0/{Idc}}} \right)} +} \\{{{Vdc} \cdot {\cos\left( {2\;\omega\; t} \right)}} + {\left( {{il}/{Idc}} \right) \cdot {Vrec}}} \\{= {{{Vrec} \cdot {drec}} + {{Vdc}\left\lbrack {{\cos^{2}\left( {\omega\; t} \right)} -} \right.}}} \\{\left. {\sin^{2}\left( {\omega\; t} \right)} \right\rbrack +} \\{\frac{{Vm} \cdot {{\sin\left( {\omega\; t} \right)}} \cdot \left( {{Im}/2} \right) \cdot {{\sin\left( {\omega\; t} \right)}}}{Idc}} \\{= {{{Vrec} \cdot {drec}} + {{Vdc} \cdot {\cos^{2}\left( {\omega\; t} \right)}}}} \\{= {Vdc}}\end{matrix} & (19)\end{matrix}$

This equation matches Equation (4). It is understood that the adoptionof the rectification duty drec′ and the discharge duty dc′ enablescontrol to make the DC voltage Vdc constant.

The corrections expressed by Equations (13) and (18) are realized by theduty correcting unit 1023. The corrections require the DC voltage Vdc,which is calculated from the original rectification duty drec and theoriginal discharge duty dc, the amplitude Vm and the phase θ, and thecommand value Vc*, as expressed by Equation (9). Alternatively, thecommand value Vdc* may be used.

Thus, it can be considered that the corrections expressed by Equations(13) and (18) are performed on the basis of the ratio (il/Idc).Furthermore, the amplitude Im of the input current is unnecessary notonly for generating the original rectification duty drec and theoriginal discharge duty dc but also for performing correction to obtainthe rectification duty drec′ and the discharge duty dc′.

E. Obtainment of DC Current Idc

As described above, the DC current Idc can be calculated from ameasurement value of the current Ish. For preparation for thedescription thereof, the following quantities are introduced. A periodT0 is one period of a carrier CA. Time periods τ4 and τ6 respectivelyrepresent the lengths of time periods over which a first state and asecond state are realized in the period T0.

The first state and the second state will be described while focusingattention on a period in which the switching element Swp is continuouslyin an OFF state and the switching element Swn is continuously in an ONstate in the inverter 5 in one period of the carrier CA. The first stateis a state in which the switching elements Sup and Svn are in an ONstate and the switching elements Sun and Svp are in an OFF state. Thesecond state is a state in which the switching elements Sup and Svp arein an ON state and the switching elements Sun and Svn are in an OFFstate. It is known that Equation (20) holds when the phase θv of thevoltage output from the inverter 5 and a coefficient of 0<k<1 areintroduced (see, for example, Japanese Patent Application Laid-Open No.2016-103961).[Math. 20]τ0/T0=1−k·sin(θv+π/3)τ4/T0=k·sin(π/3−θv)τ6/T0=k·sin(θv)  (20)

Likewise, the phase θi and amplitude Io of the load current iu areintroduced, the load currents iu, iv, and iw forming three-phase AC, andEquation (21) holds.[Math. 21]iu=Io·cos(θi)iv=Io·cos(θi−2π/3)iw=Io·cos(θi+2π/3)  (21)

The current Ia in Equation (22) is used as the DC current Idc. Theperiod T0 is one period of the carrier CA. The time periods τ4 and τ6respectively represent the lengths of time periods over which the firststate and the second state are realized in the period T0. CurrentsIsh(t4) and Ish(t6) respectively represent the values of the current Ishmeasured at time t4 and time t6. Time t4 and time t6 are respectivelyselected from the time points at which the first state and the secondstate are realized.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 22} \right\rbrack & \; \\{{Ia} = \frac{{\tau\;{4 \cdot {{Ish}\left( {t\; 4} \right)}}} + {\tau\;{6 \cdot {{Ish}\left( {t\; 6} \right)}}}}{To}} & (22)\end{matrix}$

Thus, in the first state, the load current iu flows from the outputterminal Pu to the inductive load 6 and flows from the output terminalsPv and Pw to the DC power supply line LL. In the second state, the loadcurrents iu and iv flow from the output terminals Pu and Pv,respectively, to the inductive load 6, and flow from the output terminalPw to the DC power supply line LL. On the basis of the relationshipiu+iv+iw=0, Equation (23) is obtained.[Math. 23]Ish(t4)=iu,Ish(t6)=−iw  (23)

Equation (22) is deformed into the following Equation (24) by usingEquations (20), (21), and (23). Here, ψ=θv−θi is the phase differencebetween the voltage and current output from the inverter 5, and thus cosψ represents the power factor of the inverter 5.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 24} \right\rbrack & \; \\\begin{matrix}{{Ia} = {k \cdot {Io} \cdot \left\lbrack {{{\sin\left( {{\pi/3} - {\theta\; v}} \right)} \cdot {\cos\left( {\theta\; o} \right)}} - {{\sin\left( {\theta\; v} \right)} \cdot {\cos\left( {{\theta\; i} + {2{\pi/3}}} \right)}}} \right\rbrack}} \\{= {\frac{k \cdot {Io}}{2}\left\lbrack \left\{ {{{\sin\left( {\pi/3} \right)} \cdot {\cos\left( {{\theta\; v} - {\theta\; i}} \right)}} - {{\cos\left( {\pi/3} \right)} \cdot {\sin\left( {{\theta\; v} - {\theta\; i}} \right)}} +} \right. \right.}} \\{\left. {{{\sin\left( {\pi/3} \right)} \cdot {\cos\left( {{\theta\; v} + {\theta\; i}} \right)}} - {{\cos\left( {\pi/3} \right)} \cdot {\sin\left( {{\theta\; v} + {\theta\; i}} \right)}}} \right\} -} \\{\left\{ {{{\sin\left( {2\;{\pi/3}} \right)} \cdot {\cos\left( {{\theta\; v} + {\theta\; i}} \right)}} + {{\cos\left( {2{\pi/3}} \right)} \cdot {\sin\left( {{\theta\; v} + {\theta\; i}} \right)}} +} \right.} \\{{{\sin\left( {{- 2}\;{\pi/3}} \right)} \cdot {\cos\left( {{\theta\; v} - {\theta\; i}} \right)}} +} \\\left. \left. {{\cos\left( {{- 2}\;{\pi/3}} \right)} \cdot {\sin\left( {{\theta\; v} - {\theta\; i}} \right)}} \right\} \right\rbrack \\{= {\frac{k \cdot {Io}}{2}\left\lbrack {{{\cos\left( {{\theta\; v} - {\theta\; i}} \right)}\left\{ {{\sin\left( {\pi/3} \right)} - {\sin\left( {{- 2}\;{\pi/3}} \right)}} \right\}} +} \right.}} \\{{{\cos\left( {{\theta\; v} + {\theta\; i}} \right)}\left\{ {{\sin\left( {\pi/3} \right)} - {\sin\left( {2\;{\pi/3}} \right)}} \right\}} +} \\{{{\sin\left( {{\theta\; v} - {\theta\; i}} \right)}\left\{ {{- {\cos\left( {\pi/3} \right)}} - {\cos\left( {{- 2}\;{\pi/3}} \right)}} \right\}} +} \\\left. {{\sin\left( {{\theta\; v} + {\theta\; i}} \right)}\left\{ {{- {\cos\left( {\pi/3} \right)}} - {\cos\left( {2\;{\pi/3}} \right)}} \right\}} \right\rbrack \\{= {{\frac{\sqrt{3} \cdot k \cdot {Io}}{2}{\cos\left( {{\theta\; v} - {\theta\; i}} \right)}} = {\frac{\sqrt{3} \cdot k \cdot {Io}}{2}\cos\;\psi}}}\end{matrix} & (24)\end{matrix}$

On the other hand, Equation (25) holds when the power factor cos ψ isused. A voltage Vr and a current il are effective values of the voltageand current, respectively, output from the inverter 5.[Math. 25]√{square root over (3)}·Vr·ir·cos ψ=Vdc·Idc  (25)(where Vr=k·Vdc/√{square root over (2)},ir=Io/√{square root over (2)})

The following Equation (26) is obtained from Equation (25).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 26} \right\rbrack & \; \\\begin{matrix}{{Idc} = {{\left( {\sqrt{3} \cdot k \cdot {{Vdc}/\sqrt{2}}} \right) \cdot \left( {{Io}/\sqrt{2}} \right) \cdot \cos}\;\psi}} \\{= {{\frac{\sqrt{3} \cdot k \cdot {Io}}{2} \cdot \cos}\;\psi}}\end{matrix} & (26)\end{matrix}$

From the comparison between Equations (24) and (26), it is understoodthat the current Ia can be adopted as the DC current Idc.

The calculation based on Equation (22) is executed by a link currentcalculating unit 1024 included in the discharge controller 102. Sincethe period T0 is a preset value and the time periods τ4 and τ6 can beobtained from the signal waves M, the current Ish and the signal waves Mare input to the link current calculating unit 1024, and the DC currentIdc is calculated.

F. Example of Reactor Current il

In the above-described manner, buffering of power can be performed whilemaking the input current Iin sinusoidal, even if the waveform of thereactor current il does not exhibit the absolute value of a sine wave.Thus, hereinafter a description will be given of an operation accordingto the present embodiment by using as an example the reactor current ilwhen simple switching is adopted. Flow of the reactor current il causedby simple switching reduces noise generated by the switch Sl, and thereduction of distortion of the input current Iin from a sinusoidalwaveform can reduce power supply harmonics.

FIG. 7 is a graph showing the waveform of the reactor current il whensimple switching is adopted. Here, a region where the phase ωt is 0 to180 degrees is shown. The waveform of the value il0 is also shown forreference.

In simple switching, the switch Sl is brought into conduction at a phaseof 0 degrees, the switch Sl is brought out of conduction at the phaseωt=ϕ(0<ϕ<180) (degrees), and the non-conduction state of the switch Slis maintained until the phase reaches 180 degrees. That is, in simpleswitching, the switch Sl is turned on once and turned off once duringone period of the rectified voltage Vrec (this is equal to a half-periodof the single-phase AC voltage Vin).

The phase ϕ is set in the following manner, for example. In Uesgi andfour others, “Single-Phase Twice voltage PFC Converter for airconditioner”, The transactions of the Institute of Electrical Engineersof Japan D, Vol. 119, No. 5, pp. 592-598, 1999, FIG. 10 illustrates therelationship among a value Ts corresponding to the time period overwhich the switch Sl is conducting, the voltage Vc across the capacitor,a power factor, and output power. For example, the inductance of thereactor L4 is 1 H, Vm=1 V, Vc=1.14 V, and 45 mW is adopted as the powerthat makes the power factor maximum. At this time, Ts=0.194. Thiscorresponds to a phase of ϕ≈70 (degrees). Thus, in the graph shown inFIG. 7, the reactor current il increases when the phase ωt is 0 to 70degrees. The reactor current il decreases when the phase ωt is 70 to 180degrees, and keeps a value of 0 after reaching 0 even if the phase ωtproceeds.

As will be described below, the phase at which the reactor current ildecreases to reach a value of 0 is about 170 degrees. Likewise, in therange where the phase ωt is 180 degrees to 360 degrees, the reactorcurrent il increases to have a maximum value during a time period overwhich the phase ωt is 180 degrees to about 250 (=180+70) degrees, anddecreases to reach a value of 0 when the phase ωt is about 250 to about350 (=170+180) degrees.

When Vm=1 and Im=√2 adopted in the above description are used, the poweris calculated to be 1/2√2. Thus, the inductance of the reactor L4 is(0.045×1)/(1/2√2)=0.127 (H) when it is calculated on the basis of Suga,Kimata, Uchida, “A Simple Switching Method for A Improved Power FactorType Single Phase Converter”, The transactions of the Institute ofElectrical Engineers of Japan D, Vol. 116, No. 4, pp. 420-426, 1996,with the frequency of the single-phase AC voltage Vin being 1 Hz.

In accordance with Suga, Kimata, Uchida, “A Simple Switching Method forA Improved Power Factor Type Single Phase Converter”, The transactionsof the Institute of Electrical Engineers of Japan D, Vol. 116, No. 4,pp. 420-426, 1996, a more specific waveform of the reactor current il iscalculated. In a time period over which the switch Sl is conducting,il=Ip·(1−cos(ωt)). Because Vm=1, Ip=1/(2π)/0.127=1.253. Thus, in FIG. 7,the local maximum value of the reactor current il is1.253×(1−cos(ϕ)=0.82 (ϕ)=70 (degrees)).

When il>0 in the time period over which the switch Sl is not conducting,il=Ip·(1−cos(ωt))−(Vc/L)·(ωt−ϕ)/ω. Here, the value L is the inductanceof the reactor L4. In the above-described example,Vc/L=1.14/0.127=8.976.

G. Behavior of Quantities

FIG. 8 includes graphs showing an operation of the direct powerconverter illustrated in FIG. 1, and illustrates an operation in a casewhere the duties drec′, dc′, and dz′ (=1−drec′−dc′) are set on the basisof the present embodiment. The waveform illustrated in FIG. 7 is adoptedas the waveform of the reactor current il.

Also in FIG. 8, individual values are converted with the amplitude Imbeing √2 and the amplitude Vm being 1, Vc=1.14 Vm and Vdc=0.86 Vm areset. Thus, these conditions are the same as in FIG. 6. FIG. 8illustrates the quantities similarly to FIG. 6. Specifically, the topgraph shows the duties drec′, dc′, and dz′; the second graph from thetop shows the DC voltage Vdc, the voltages drec′·Vrec and dc′·Vc (seeEquation (19)) that constitute the DC voltage Vdc, and the DC currentIdc; the third graph from the top shows the currents irec, ic, il, andirec1; and the bottom graph shows the instantaneous powers Pin, Pdc,Pbuf, Pc, −Pl, and Prec.

When simple switching is adopted to cause the reactor current il to flowthrough the power buffer circuit 4, control using the rectification dutydrec′ and the discharge duty dc′ makes not only the inverter input powerPdc but also the DC voltage Vdc and the DC current Idc constant, andalso causes the waveform of the current irec to exhibit the absolutevalue of a sine wave. Accordingly, it is understood that the waveform ofthe input current Iin is sinusoidal.

In addition, it is understood from the waveforms of the instantaneouspowers Pin, Pdc, and Pbuf that the power buffer circuit 4 functions as apower buffer.

Note that, in the case illustrated in FIG. 8, there is a phase regionwhere dz′<0. In this region, the rectification duty drec′ and the zeroduty dz′ are corrected to achieve drec′=1−dc′ and dz′=0. The correctionof the rectification duty for obtaining a non-negative zero duty in allthe phases is known from, for example, Japanese Patent No. 5794273, andthus the details thereof is omitted here.

In FIG. 9, individual values are converted with the amplitude Im being√2 and the amplitude Vm being 1, Vc=1.14 Vm is set. Note that Equation(19) does not hold in the region where drec′=1−dc′. Thus, in the regionwhere dz′=0, the DC voltage Vdc is under 0.86 Vm, and the minimum valueof the DC voltage Vdc is 0.80 Vm. Accordingly, it is understood that thewaveform of the current irec is slightly distorted from the absolutevalue of a sine wave in the region where dz′=0. Thus, it is understoodthat, although the inverter input power Pdc is also slightly distortedfrom a constant value relative to the phase, the power buffer circuit 4functions as a power buffer.

When such distortion is beyond the allowance of distortion of the inputcurrent Iin from a sinusoidal waveform, the zero duty can be madenon-negative by using another method.

FIG. 10 includes graphs showing an operation of the direct powerconverter illustrated in FIG. 1, and illustrates an operation in a casewhere the duties drec′, dc′, and dz′ (=1−drec′−dc′) are set on the basisof the present embodiment. The waveform illustrated in FIG. 7 is adoptedas the waveform of the reactor current il. The quantities areillustrated in FIG. 10 similarly to FIG. 8.

Also in FIG. 10, individual values are converted with the amplitude Imbeing √2 and the amplitude Vm being 1. Note that, although Vc=1.14 Vm,the DC voltage Vdc is lower than in FIG. 8 and FIG. 9, that is, Vdc=0.76Vm.

Such a lower DC voltage Vdc makes the original rectification duty drecand the original discharge duty dc lower (see Equations (7) and (8)) andmakes the rectification duty drec′ and the discharge duty dc′ loweraccordingly, and the minimum value of dz′ can be 0 in all the phases inthe control using these duties.

Increasing the command value Vc* with reference to Equation (8)(performing control to increase the voltage Vc across the capacitor C4)makes the original discharge duty dc lower (see Equation (8)) and makesthe discharge duty dc′ lower accordingly, and dz′≥0 can be satisfied inall the phases in the control using these duties. Note that, in thiscase, the quantities described above in “F. Example of reactor currentil” are changed, or the power factor of the power buffer circuit 4fluctuates.

H. Decrease in Size of Reactor and Improvement of DC Voltage Vdc

(h-1) Relationship Between Boost Operation and Inductance

When simple switching is adopted in the present embodiment, theinductance required for the reactor increases compared with thetechnique exemplified in Uesgi and four others, “Single-Phase Twicevoltage PFC Converter for air conditioner”, The transactions of theInstitute of Electrical Engineers of Japan D, Vol. 119, No. 5, pp.592-598, 1999. This is because, in the present embodiment, unlike in thetechnique exemplified in Uesgi and four others, “Single-Phase Twicevoltage PFC Converter for air conditioner”, The transactions of theInstitute of Electrical Engineers of Japan D, Vol. 119, No. 5, pp.592-598, 1999, a boost operation is performed in simple switching.

Uesgi and four others, “Single-Phase Twice voltage PFC Converter for airconditioner”, The transactions of the Institute of Electrical Engineersof Japan D, Vol. 119, No. 5, pp. 592-598, 1999 exemplifies the case ofobtaining a DC voltage of 270 V from a power supply of 100 V (aneffective voltage value of 100 V in single phase) by usingvoltage-doubling rectification. When the inductance is converted to 1 Hand the peak value of the voltage corresponding to the amplitude Vm ofthe present embodiment is converted to 1 V, Vc=270/2/(100×√2)≈0.95. Inaddition, Uesgi and four others, “Single-Phase Twice voltage PFCConverter for air conditioner”, The transactions of the Institute ofElectrical Engineers of Japan D, Vol. 119, No. 5, pp. 592-598, 1999exemplifies a case where the input power is 1800 W and where the timeperiod over which the switch is conducting is 2.8 ms in a switchingperiod of 50 Hz. In this case, Ts=(2.8/1000)×50=0.14. At this time, inview of the graph illustrated in FIG. 10 of Uesgi and four others,“Single-Phase Twice voltage PFC Converter for air conditioner”, Thetransactions of the Institute of Electrical Engineers of Japan D, Vol.119, No. 5, pp. 592-598, 1999, the power that makes the power factormaximum is about 30 mW. Compared with this power, the power (45 mW)exemplified in the description using FIG. 7 is high. This is because, inthe present embodiment, Vc=1.14 (>0.96) is obtained through a boostoperation.

According to the equation for calculating an inductance shown in Uesgiand four others, “Single-Phase Twice voltage PFC Converter for airconditioner”, The transactions of the Institute of Electrical Engineersof Japan D, Vol. 119, No. 5, pp. 592-598, 1999, the value of theinductance is calculated as follows: 30 [mW]×(100 [V]×√2)²/1800 [W]/50[Hz]=6.7 [mH]. On the other hand, in a circuit constant table, the valueof the inductance is 6.2 mH. The value of the inductance necessary forvoltage-doubling rectification exemplified in Uesgi and four others,“Single-Phase Twice voltage PFC Converter for air conditioner”, Thetransactions of the Institute of Electrical Engineers of Japan D, Vol.119, No. 5, pp. 592-598, 1999 is assumed to be about 6.5 mH. This isconverted to 6.5 [mH]×(200/100)²=26 [mH] when a power supply of 200 V isused.

In contrast to Uesgi and four others, “Single-Phase Twice voltage PFCConverter for air conditioner”, The transactions of the Institute ofElectrical Engineers of Japan D, Vol. 119, No. 5, pp. 592-598, 1999, thepower is 45/30 times in the present embodiment. When the single-phase ACpower supply 1 according to the present embodiment is a power supply of200 V, Vm=230 V. Thus, compared with voltage-doubling rectificationadopted in Uesgi and four others, “Single-Phase Twice voltage PFCConverter for air conditioner”, The transactions of the Institute ofElectrical Engineers of Japan D, Vol. 119, No. 5, pp. 592-598, 1999, aninfluence of full-wave rectification adopted in the present embodimenton the inductance is (230/100)² times. Thus, a necessary value of theinductance is 6.5 [mH]×45/30×(230/100)²≈52 [mH].

Accordingly, the adoption of the simple switching exemplified in Uesgiand four others, “Single-Phase Twice voltage PFC Converter for airconditioner”, The transactions of the Institute of Electrical Engineersof Japan D, Vol. 119, No. 5, pp. 592-598, 1999 in the power buffercircuit 4 in which a boost operation is performed is likely to increasethe inductance of the reactor L4 (about twice in the above example).

(h-2) Relationship Between Decrease in Inductance and DC Voltage Vdc

Suga, Kimata, Uchida, “A Simple Switching Method for A Improved PowerFactor Type Single Phase Converter”, The transactions of the Instituteof Electrical Engineers of Japan D, Vol. 116, No. 4, pp. 420-426, 1996introduces a technique for delaying the starting time of conduction of aswitching element. According to this technique, a large power factor canbe obtained even when the inductance is small.

FIG. 11 is a graph showing the relationship between an inductance L anda power factor when a power of 1.84 kW (an effective voltage value of230 V and an effective current value of 8 A) is adopted in the presentembodiment on the basis of Suga, Kimata, Uchida, “A Simple SwitchingMethod for A Improved Power Factor Type Single Phase Converter”, Thetransactions of the Institute of Electrical Engineers of Japan D, Vol.116, No. 4, pp. 420-426, 1996. Note that the frequency of thesingle-phase AC voltage Vin is 50 Hz. Graphs G0, G1, G2, and G3respectively show the cases where the phase at which the switch Sl isbrought into conduction (hereinafter referred to as a “conduction startphase”) is 0 degrees, 27 (=360×0.075) degrees, 36 (=360×0.1) degrees,and 45 (=360×0.125) degrees relative to a reference phase at which therectified voltage Vrec is 0.

It is understood from these graphs that the inductance L has a valuethat makes the power factor maximum. In each of the graphs G0, G1, G2,and G3, the data that makes the power factor maximum is plotted inwhite.

The power factor decreases as the conduction start phase increases.However, the value of the inductance L that makes the power factormaximum decreases as the conduction start phase increases.

For example, in the graph G0, the power factor is maximum when theinductance L is 51.37 mH. In the cases illustrated in FIG. 8, FIG. 9,and FIG. 10, under the conditions where the frequency is 50 Hz, theeffective voltage value is 230 V, and the effective current value is 8A, the inductance L that makes the power factor maximum is 51.75 mH(=0.045×(230×√2)²/(1840×50) (H)), which is almost equal to a value ofthe inductances L that make the power factor maximum.

When the direct power converter according to the present embodiment isadopted to a compact air conditioner used with a power of, for example,2 kW or less, it is desired that the inductance of the reactor L4 be setto 15 mH to 30 mH and a compact component be adopted accordingly.

For example, when an inductance L of about 28 mH is adopted, it isdesired that the conduction start phase be 45 degrees (see the graph G3)from the viewpoint of a flat power factor characteristic. However, fromthe viewpoint of increasing the power factor, it is desired that theconduction start phase be 36 degrees (see the graph G2). In thedescription given below, the conduction start phase is 36 degrees in allthe cases.

FIG. 12 to FIG. 14 each include graphs showing an operation of thedirect power converter illustrated in FIG. 1. In these figures, as inFIG. 8 to FIG. 11, graphs are drawn with Vc being 1.14 Vm, the amplitudeIm being √2, and the amplitude Vm being 1.

A power of 1.84 kW (an effective voltage value of 230 V and an effectivecurrent value of 8 A) is adopted, and the inductance L is set to 28.54mH. The conduction start phase is 36 degrees, and thus the phase atwhich the switch Sl shifts from a conduction state to a non-conductionstate is about 68 (360×0.19) degrees. At this phase, the reactor currentil has a maximum value. After that, the reactor current il decreases asthe phase increases, and the reactor current il becomes 0 at about 155(360×0.430) degrees.

In the present embodiment, full-wave rectification is performed, andthus the switch Sl is conducting when the phase is in the range fromabout 216 (=36+180) degrees to about 248 (=68+180) degrees, and thereactor current il flows when the phase is in the range from about 216degrees to about 335 (=155+180) degrees and has a maximum value evenwhen the phase is about 248 (=68+180) degrees.

The cases illustrated in FIG. 8 to FIG. 11 correspond to a case where apower of 1.84 kW (an effective voltage value of 230 V and an effectivecurrent value of 8 A) is adopted and the inductance L is set to 51.37mH. The reactor current il flows when the phase is in the range from 0degrees to about 170 (=360×0.471) degrees and in the range from 180degrees to about 350 (=170+180) degrees, and has a maximum value whenthe phase is about 70 (=360×0.194) degrees and about 250 (=70+180)degrees.

FIG. 12 illustrates an operation in a case where the rectification dutydrec′ is set on the basis of Equation (13), the discharge duty dc′ isset on the basis of Equation (18), the zero duty dz′ is set by1−drec′−dc′, and Vdc is set to 0.86 Vm, as in the conditions used inFIG. 8.

FIG. 12 illustrates quantities similarly to FIG. 8. Specifically, thetop graph shows the duties drec′, dc′, and dz′; the second graph fromthe top shows the DC voltage Vdc, the voltages drec′·Vrec and dc′·Vcthat constitute the DC voltage Vdc, and the DC current Idc; the thirdgraph from the top shows the currents irec, ic, il, and irec1; and thebottom graph shows the instantaneous powers Pin, Pdc, Pbuf, Pc, −Pl, andPrec.

As in the description using FIG. 8, it is understood that there is aphase region where dz′<0 with the waveform of the input current Iinbeing sinusoidal. Furthermore, in this region, the absolute value of thezero duty dz′ is larger in the case illustrated in FIG. 12 than in thecase illustrated in FIG. 8.

FIG. 13 illustrates quantities in a case where drec′=1−dc′ and dz′=0 isadopted in the region where 1−drec′−dc′<0, as in the conditions used inFIG. 9. FIG. 13 illustrates quantities similarly to FIG. 12. As in thedescription using FIG. 9, in the region where dz′=0, the DC voltage Vdcis under 0.86 Vm, and the waveform of the current irec is distorted fromthe absolute value of a sine wave. The minimum value of the DC voltageVdc is 0.72 Vm in FIG. 13, which is lower than 0.80 Vm, which is theminimum value in FIG. 9.

FIG. 14 illustrates quantities in a case where the DC voltage Vdc is setso that there is not a phase region where dz′=1−drec′−dc′<0, with the DCvoltage Vdc being constant, and corresponds to FIG. 10. Under theconditions used in FIG. 10, dz′≥0 can be satisfied when Vdc=0.76 Vm.Under the conditions used in FIG. 14, the inductance L is 28.54 mH andthe conduction start phase is 36 degrees, and thus it is necessary todecrease Vdc to 0.69 Vm to satisfy 1−drec′−dc′≥0.

The cause of a decrease in the minimum value of the DC voltage Vdc thatfluctuates and a decrease in the maximum value of the DC voltage Vdcthat can be constant may be that a decrease in the inductance L causes ashorter time period over which the reactor current il flows and that thereactor current il contains many harmonic components (specifically, thethird- or higher-order odd harmonic components of the frequency of thesingle-phase AC voltage Vin).

(h-3) Control on Harmonics and Improvement of DC Voltage Vdc

Harmonics are controlled by, for example, the standard IEC 61000-3-2.For example, in the reactor current il exemplified in FIG. 12 to FIG.14, the magnitude of a third-order harmonic component is larger thanthat in the standard IEC 61000-3-2. Since irec=irec1+il (see FIG. 1 andFIG. 4), the third-order harmonic component of the reactor current il isreflected as is in the current irec and affects the third-order harmoniccomponent of the input current Iin accordingly. Hereinafter, a techniquefor reducing harmonics contained in the reactor current il will beconsidered.

FIG. 15 is a graph showing the waveform of the reactor current ilexemplified in FIG. 12 to FIG. 14 in the half-period of the single-phaseAC voltage Vin, specifically, when the phase ωt is 0 to 180 degrees.Also when the phase ωt is 180 to 360 degrees, the reactor current il hasthe same waveform. As in the above case, a power of 1.84 kW (aneffective voltage value of 230 V and an effective current value of 8 A)is adopted, and the inductance L is set to 28.54 mH. The reactor currentil is shown with the amplitude Im being √2.

For example, in the standard IEC 61000-3-2, a third-order harmoniccomponent of up to an effective value of 2.30 A is allowed under thecondition of a rated voltage of 230 V. On the other hand, the reactorcurrent il illustrated in FIG. 15 contains a third-order harmoniccomponent of an effective value of 2.87 A when the effective voltagevalue is 230 V, and harmonic components of other orders are within anallowable range of the standard IEC 61000-3-2. Thus, to reduce thethird-order harmonic component of 0.571 (=2.87−2.30) A, which issubtracted from the reactor current il as the amount of correction.

FIG. 16 is a graph showing the waveform of the input current Iin whenthe amount of correction is subtracted from the reactor current il asdescribed above. Note that Vc=1.14 Vm, the amplitude Vm is 1, and theamplitude Im of the input current Iin before subtraction of the amountof correction is √2. As a result of subtracting the amount ofcorrection, the harmonics of the input current Iin become within theallowable range of the standard IEC 61000-3-2.

To perform such subtraction of the amount of correction, a rectificationduty drec″, a discharge duty dc″, and a zero duty dz″ are set in thefollowing manner.

First, an amount of correction il′ of the reactor current il isintroduced. In the example given above, the amount of correction il′ isthe amount of reduction (0.571 A) in the third-order harmonic componentof the reactor current il. In Equation (12), the amount of deviation ofthe reactor current il from the value il0, (il−i0) is used. Thus, tocalculate the rectification duty drec″ from the original rectificationduty drec by subtracting the amount of correction il′, the rectificationduty drec′ may be replaced with the rectification duty drec″ and theamount of deviation (il−i0) may be replaced with a value (−il′) informally in Equation (12). Furthermore, Equation (27) is obtained inview of Equation (7).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 27} \right\rbrack & \; \\\begin{matrix}{{drec}^{''} = {{drec} + \left( \frac{{il}^{\prime}}{Idc} \right)}} \\{= {{\frac{Vdc}{Vm} \cdot {{\sin\left( {\omega\; t} \right)}}} + \left( \frac{{il}^{\prime}}{Idc} \right)}}\end{matrix} & (27)\end{matrix}$

Likewise, the discharge duty dc′ is replaced with the discharge duty dc″and the amount of deviation (il−i0) is replaced with the value (−il′)formally in Equation (14), so that Equation (28) is obtained.Furthermore, Equation (29) is obtained in view of Equation (8).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 28} \right\rbrack & \; \\{{{Vc}*{\cdot {Idc} \cdot \left( {{dc}^{''} - {dc}} \right)}} = {{- {il}^{\prime}} \cdot {Vm} \cdot {{\sin\left( {\omega\; t} \right)}}}} & (28) \\\left\lbrack {{Math}.\mspace{14mu} 29} \right\rbrack & \; \\\begin{matrix}{{dc}^{''} = {{dc} - \frac{\left( \frac{{il}^{\prime}}{Idc} \right) \cdot {Vm} \cdot {{\sin\left( {\omega\; t} \right)}}}{{Vc}*}}} \\{= {{\frac{Vdc}{{Vc}*} \cdot {\cos^{2}\left( {\omega\; t} \right)}} - {\left( \frac{{il}^{\prime}}{Idc} \right) \cdot \left( \frac{Vm}{{Vc}*} \right) \cdot {{\sin\left( {\omega\; t} \right)}}}}}\end{matrix} & (29)\end{matrix}$

Calculation is performed similarly to Equation (19) by using therectification duty drec″ and the discharge duty dc″ obtained in thismanner, so that Equation (30) is obtained. That is, it is understoodthat the adoption of the rectification duty drec″ and the discharge dutydc″ enables control for making the DC voltage Vdc constant.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 30} \right\rbrack & \; \\\begin{matrix}{{{{Vrec} \cdot {drec}^{''}} + {{Vc}*{\cdot {dc}^{''}}}} = {{{Vrec} \cdot \left\lbrack {{drec} + {{il}^{\prime}/{Idc}}} \right\rbrack} +}} \\{{{Vdc} \cdot {\cos^{2}\left( {\omega\; t} \right)}} + {\left( {{il}^{\prime}/{Idc}} \right) \cdot}} \\{{Vm} \cdot {{\sin\left( {\omega\; t} \right)}}} \\{= {{{Vrec} \cdot {drec}} + {{Vrec} \cdot \left( {{il}^{\prime}/{Idc}} \right)} +}} \\{{{Vdc} \cdot {\cos^{2}\left( {\omega\; t} \right)}} - {\left( {{il}^{\prime}/{Idc}} \right) \cdot {Vrec}}} \\{= {{{Vrec} \cdot {drec}} + {{Vdc} \cdot {\cos^{2}\left( {\omega\; t} \right)}} +}} \\{\left( \frac{{il}^{\prime}}{Idc} \right)\left\lbrack {{Vrec} - {{Vm} \cdot {{\sin\left( {\omega\; t} \right)}}}} \right\rbrack} \\{= {{{Vrec} \cdot {drec}} + {{Vdc} \cdot {\cos^{2}\left( {\omega\; t} \right)}}}} \\{= {Vdc}}\end{matrix} & (30)\end{matrix}$

Note that, as described using FIG. 8 and FIG. 12, when the zero duty dz″is set by 1−drec″−dc″, there is a phase region where dz″<0. FIG. 17includes graphs showing quantities in a case where the rectificationduty drec″ and the discharge duty dc″ are set by Equations (27) and(29), respectively, and the zero duty dz″ is set by 1−drec″−dc″, withthe values of specifications corresponding to those in the descriptionusing FIG. 12. It is understood that there is a phase region wheredz″<0.

FIG. 18 includes graphs showing quantities in a case where therectification duty drec″ and the discharge duty dc″ are set by Equations(27) and (29), respectively, the zero duty dz″ is set by 1−drec″−dc″,and modification is performed to satisfy dz″=0 and drec″=1−dc″ only in aphase where 1−drec″−dc″<0, with the values of specificationscorresponding to those in the description using FIG. 13.

The zero duty dz″ illustrated in FIG. 17 is smaller in the absolutevalue of a negative value than the zero duty dz′ illustrated in FIG. 12.Thus, the minimum value of the DC voltage Vdc is larger in the caseillustrated in FIG. 18 than in the case illustrated in FIG. 13.Specifically, in the region where dz″>0, the DC voltage Vdc is constantat 0.86 Vm, whereas when dz″=0 the minimum value of the DC voltage Vdcis 0.84 Vm (in the case illustrated in FIG. 13, the minimum value of theDC voltage Vdc is 0.72 Vm).

The DC voltage Vdc can be kept small and constant as in FIG. 14 whilemaintaining the rectification duty drec″, the discharge duty dc″, anddz″=1−drec″−dc″ in all the phases. FIG. 19 includes graphs showingquantities in a case where the rectification duty drec″, the dischargeduty dc″, and the zero duty dz″ are set similarly to FIG. 17. Setting ofVdc=0.82 Vm enables the minimum value of the zero duty dz″ to be 0, nota positive value. That is, the maximum value of the DC voltage Vdc thatcan be kept constant is larger than in the case illustrated in FIG. 14(Vdc=0.69 Vm).

As a result of decreasing the inductance L to reduce the size of thereactor L4 and suppressing harmonic components of the reactor current ilgenerated accordingly in the manner described above, the DC voltage Vdcthat can be used for power conversion by the inverter 5 is increased,with the harmonics of the input current Iin being reduced and distortionof the waveform of the input current Iin from a sinusoidal waveformbeing allowed.

A decrease in the inductance L causes an increase in components of thefifth- or higher-order harmonics of the reactor current il. Thus, it maybe desirable to adopt other harmonic components as the amount ofcorrection il′.

FIG. 20 is a graph showing the waveform of the reactor current il in thehalf-period of the single-phase AC voltage Vin, specifically, when thephase ωt is 0 to 180 degrees. Also when the phase ωt is 180 to 360degrees, the reactor current il has the same waveform.

Here, as in the case described using FIG. 15, a power of 1.84 kW (aneffective voltage value of 230 V and an effective current value of 8 A)is adopted, but the inductance L is 18 mH, which is smaller than in thecase described using FIG. 15. The amplitude of the reactor current il isconverted with the amplitude Im of the input current Iin being √2.

Since the conduction start phase is 36 degrees, the phase at which theswitch Sl shifts from a conduction state to non-conduction state isabout 61 (360×0.17) degrees. At this phase, the reactor current il has amaximum value. At about 141 (360×0.394) degrees, the reactor current ilis 0.

For example, in the standard IEC 61000-3-2, a third-order harmoniccomponent, a seventh-order harmonic component, and an eleventh-orderharmonic component are allowed up to effective values of 2.30 A, 0.77 A,and 0.33 A, respectively, under the condition of a rated voltage of 230V. On the other hand, when the effective voltage value is 230 V, thereactor current il illustrated in FIG. 20 contains a third-orderharmonic component of an effective value of 3.66 A, a seventh-orderharmonic component of an effective value of 1.17 A, and aneleventh-order harmonic component of an effective value of 0.38 A, andharmonic components of other orders are within an allowable range of thestandard IEC 61000-3-2.

Thus, as the amount of correction il′, the sum of the third-orderharmonic component of 1.36 (=3.66−2.30) A, the seventh-order harmoniccomponent of 0.40 (=1.17−0.77) A, and the eleventh-order harmoniccomponent of 0.05 (=0.38−0.33) A is adopted.

FIG. 21 is a graph showing the waveform of the input current Iin whenthe amount of correction il′ is subtracted from the reactor current ilas described above. Note that Vc=1.14 Vm, the amplitude Vm is 1, and theamplitude Im of the input current Iin before subtraction of the amountof correction il′ is √2. As a result of subtracting the amount ofcorrection il′, the harmonic components of the input current Iin becomewithin the allowable range of the standard IEC 61000-3-2.

FIG. 22 includes graphs showing quantities in a case where therectification duty drec″ and the discharge duty dc″ are set by Equations(27) and (29), respectively, the zero duty dz″ is set by 1−drec″−dc″,and modification is performed to satisfy dz″=0 and drec″=1−dc″ only in aphase where 1−drec″−dc″<0, with the values of specifications other thanthe inductance L corresponding to those in the description using FIG.18.

In the region where dz″>0, the DC voltage Vdc is constant at 0.86 Vm,whereas when dz″=0 the minimum value of the DC voltage Vdc is 0.81 Vm.This is lower than the minimum value 0.84 Vm when the inductance L is 28mH (in the case described using FIG. 18).

FIG. 23 includes graphs showing quantities in a case where Vdc=0.78 Vm.Such a setting enables the minimum value of the zero duty dz″ to be 0,not a positive value. That is, even if the inductance L is small atabout 18 mH, the adoption of the rectification duty drec″, the dischargeduty dc″, and the zero duty dz″ enables the maximum value of the DCvoltage Vdc that is kept constant to be larger than in a case where theinductance L is about 51 mH, and the rectification duty drec′, thedischarge duty dc′, and the zero duty dz′ are adopted (see FIG. 10:Vdc=0.76 Vm). Furthermore, the harmonic components of the input currentIin can be within the allowable range of the standard.

On the basis of the similarity between Equations (12) and (27) and inview of Equation (7), the rectification duties drec′ and drec″ can bedefined as follows:

the rectification duties (drec′, drec″) are set on the basis of theproduct (Vdc/Vm)·|sin(ωt)| of the ratio Vdc/Vm of the DC voltage Vdc tothe amplitude Vm and the absolute value |sin(ωt)| of the sinusoidalvalue of the phase cot; and the ratio (il/Idc, il′/Idc) of the “current”(il, il′) to the DC current Idc.

The foregoing product represents the original rectification duty drec(see Equation (7)). The foregoing “current” is the reactor current ilfor the rectification duty drec′ and is the amount of correction il′ forthe rectification duty drec″. The amount of correction il′ is the amountof reduction in the harmonic component of the order to be reduced.

The rectification duty drec′ has a value obtained by subtracting theratio il/Idc from twice the product (Vdc/Vm)·|sin(ωt)| (see Equation(13)). The rectification duty drec″ has a value obtained by adding theratio il′/Idc to the product (Vdc/Vm)·|sin(ωt)| (see Equation (27)).

The discharge duty dc′ can be expressed as follows on the basis ofEquation (16):

set on the basis of the product (Vdc/Vc)·cos(2ωt) of the ratio Vdc/Vc ofthe DC voltage Vdc to the voltage Vc across the capacitor C4 (thisaccurately follows the command value Vc*) and the cosine value cos(2ωt)of the value twice the phase cot; and the product(Vm/Vc)·(il/Idc)·|sin(ωt)| of the ratio Vm/Vc of the amplitude Vm to thevoltage Vc across the capacitor C4, the ratio il/Idc, and the absolutevalue |sin(ωt)|. Specifically, the value of the discharge duty dc′ isequal to the sum of these two products.

The discharge duty dc″ can be expressed as follows on the basis ofEquation (29):

set on the basis of a first product (Vdc/Vc)·cos²(ωt) of the ratioVdc/Vc of the DC voltage Vdc to the voltage Vc across the capacitor C4(this accurately follows the command value Vc*) and the square cos²(ωt)of the cosine value of the phase cot; and a second product(Vm/Vc)·(il′/Idc)·|sin(ωt)| of the ratio Vm/Vc of the amplitude Vm tothe voltage Vc across the capacitor C4, the ratio il′/Idc, and theabsolute value |sin(ωt)|. Specifically, the value of the discharge dutydc″ is a value obtained by subtracting the second product from the firstproduct. The first product represents the original discharge duty (seeEquation (8)).

In the derivation of each of the foregoing rectification duties drec′and drec″ and the discharge duties dc′ and dc″, the distortion of thereactor current il is not assumed to be resulted from simple switching.Thus, it is clear that the technique using the rectification dutiesdrec′ and drec″ and the discharge duties dc′ and dc″ is not assumed tobe based on simple switching.

(h-4) Exemplary Configuration for Obtaining Amount of Correction il′

FIG. 24 is a block diagram exemplifying a first configuration of thedischarge controller 102 for obtaining the rectification duty drec″ andthe discharge duty dc″ and its vicinity. The discharge controller 102constitutes the control device 10 together with the inverter controller101 and the charge controller 103 illustrated in FIG. 2.

In FIG. 24, the discharge controller 102 includes anamount-of-correction generating unit 1025 in addition to the dutycalculating unit 1021, the comparator 1022, the duty correcting unit1023, and the link current calculating unit 1024 described using FIG. 2.Note that the duty correcting unit 1023 obtains the rectification dutydrec″ and the discharge duty dc″ from the original rectification dutydrec and the original discharge duty dc, respectively, in accordancewith Equations (27) and (29) (particularly the first equality in eachequation), respectively.

FIG. 25 is a block diagram exemplifying the configuration of theamount-of-correction generating unit 1025. The amount-of-correctiongenerating unit 1025 includes an input current estimating unit 1025 aand a correction harmonic table 1025 b. The input current estimatingunit 1025 a calculates an estimated value Is of the input current Iin onthe basis of the output power Pout, the amplitude Vm, and the phase ωt.For example, the following equation can be adopted.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 31} \right\rbrack & \; \\{{Is} = \frac{2 \cdot {Pout}}{{Vm} \cdot {{\sin\left( {\omega\; t} \right)}}}} & (31)\end{matrix}$

The correction harmonic table 1025 b stores therein a table showing therelationship between the input current Iin and the amount of control ofharmonic components, searches the table while adopting the estimatedvalue Is as the input current Iin, and outputs the corresponding amountof correction il′.

That is, in the first configuration, the amount of correction il′ iscalculated on the basis of the distortion of the input current Iin.Thus, in the first configuration, the reactor current il is notnecessary for the control device 10.

FIG. 26 is a block diagram exemplifying a second configuration of thedischarge controller 102 for obtaining the rectification duty drec″ andthe discharge duty dc″ and its vicinity. The discharge controller 102also constitutes the control device 10 together with the invertercontroller 101 and the charge controller 103 illustrated in FIG. 2.

In FIG. 26, the discharge controller 102 includes anamount-of-correction generating unit 1026 in addition to the dutycalculating unit 1021, the comparator 1022, the duty correcting unit1023, and the link current calculating unit 1024 described using FIG. 2.Note that the duty correcting unit 1023 obtains the rectification dutydrec″ and the discharge duty dc″ in accordance with Equations (27) and(29) (particularly the first equality in each equation), respectively.

FIG. 27 is a block diagram exemplifying the configuration of theamount-of-correction generating unit 1026. The amount-of-correctiongenerating unit 1026 includes an FFT operation unit 1026 a, a correctionharmonic selecting unit 1026 b, and an inverse FFT operation unit 1026c. The FFT operation unit 1026 a performs fast Fourier transform on thereactor current il to calculate an amplitude Ilh(n) of each component ofthe n-th order frequency. The correction harmonic selecting unit 1026 bselects the order of harmonics exceeding the upper limit that is set foreach order. Subsequently, regarding the selected order (hereinafterreferred to as α-th order), the correction harmonic selecting unit 1026b calculates the amount of the amplitude Ilh(α) of the α-th orderharmonic component exceeding the upper limit that is set for the α-thorder harmonic, as an amount of correction Il′(α) of the α-th orderharmonic. An amount of correction Il′(β) (β≠α) of the order that is notselected is 0. The inverse FFT operation unit 1026 c performs inversefast Fourier transform by using the amount of correction Il′(n) or theamount of correction Il′(α) to calculate the amount of correction il′.

That is, in the second configuration, the amount of correction il′ iscalculated on the basis of the distortion of the reactor current il (seeFIG. 15 and FIG. 20).

I. Modification (i-1) First Modification

FIG. 28 is a circuit diagram illustrating diode bridges 3 a and 3 breplacing the converter 3 and their vicinity as a first modification ofthe power converter. This configuration is known from, for example,Yamashita, Sakakibara, “A control method of asingle-phase-to-three-phase power converter with an active buffer forincreasing voltage transfer ratio”, the Institute of ElectricalEngineers of Japan Industry Applications Society Conference 2016, 1-54,pp. I-181 to I-186, 2016.

The diode bridge 3 a includes, like the converter 3 described in theforegoing embodiment, the diodes D31, D32, D33, and D34, whichconstitute a bridge circuit. The diode bridge 3 b includes diodes D35,D36, D32, and D34, which constitute a bridge circuit. That is, the diodebridges 3 a and 3 b share the diodes D32 and D34.

In this modification, the DC power supply line LH described in theforegoing embodiment is replaced with two DC power supply lines LH1 andLH2. On the other hand, the DC power supply line LL is connected to theanodes of the diodes D32 and D34, the switch Sl on the opposite side tothe reactor L4, the capacitor C4 on the opposite side to the switch Sc,and the inverter 5, as in the foregoing embodiment.

The DC power supply line LH1 is connected to the cathodes of the diodesD31 and D33 in common, the switch Sc on the opposite side to thecapacitor C4, and the inverter 5, like the DC power supply line LH. TheDC power supply line LH2 is connected to the cathodes of the diodes D35and D36 in common, and the reactor L4 on the opposite side to the switchSl.

Thus, the supply of power to the DC power supply line LH1 from the diodebridge 3 a and the discharge circuit 4 a is equivalent to the supply ofpower to the DC power supply line LH in the foregoing embodiment fromthe converter 3 and the discharge circuit 4 a. Also, the supply of powerfrom the DC power supply line LH2 to the charge circuit 4 b via thediode bridge 3 b (the reception of power by the charge circuit 4 b) isequivalent to the supply of power from the DC power supply line LH inthe foregoing embodiment to the charge circuit 4 b via the converter 3(the reception of power by the charge circuit 4 b).

As described above, the path for supplying power and the path forreceiving power are different from each other, and thus the currentblocking circuit 4 c according to the foregoing embodiment is notnecessary in the first modification.

(i-2) Second Modification

The current blocking circuit 4 c also has a function of blocking thecurrent flowing from the discharge circuit 4 a toward the converter 3.Thus, the filter 2 can be disposed between the converter 3 and thecurrent blocking circuit 4 c.

FIG. 29 and FIG. 30 are circuit diagrams each illustrating theconfiguration on the single-phase AC power supply 1 side respect to theinverter 5 in a case where the filter 2 is disposed between theconverter 3 and the current blocking circuit 4 c in the power converter.

In the configuration illustrated in FIG. 29, the reactor L2 isinterposed between the converter 3 and the reactor L4. Thisconfiguration is known from, for example, Yamashita, Sakakibara, “Acontrol method of a single-phase-to-three-phase power converter with anactive buffer for increasing voltage transfer ratio”, the Institute ofElectrical Engineers of Japan Industry Applications Society Conference2016, 1-54, pp. I-181 to I-186, 2016.

In the configuration illustrated in FIG. 30, the reactor L4 is connectedto the converter 3-side end of the reactor L2. In other words, thefilter 2 may be included in the charge circuit 4 b although the filter 2is disposed between the converter 3 and the current blocking circuit 4c.

Specifically, the reactor L2 is provided on the DC power supply line LH,at a position across the reactor L4 from the converter 3. The capacitorC2 is connected between the DC power supply lines LH and LL, at aposition across the reactor L2 from the converter 3, and constitutes thefilter together with the reactor L2.

Accordingly, it is clear that, even when the end farther from the diodeD40 of the reactor L4 is connected between the reactor L2 and theconverter 3, harmonics resulting from the switching operation of theinverter 5 do not propagate to the single-phase AC power supply 1.

The present invention has been described in detail. The descriptiongiven above is an example in all aspects and the present invention isnot limited thereto. It is understood that an unlimited number ofmodification examples not described here may be assumed withoutdeviating from the scope of the present invention.

The invention claimed is:
 1. A control device for controlling a directpower converter, the direct power converter comprising: a DC linkincluding a first DC power supply line and a second DC power supplyline; a rectifying converter that receives a single-phase AC voltage andoutputs a ripple power to the DC link; a power buffer circuit thatsupplies and receives a power to and from the DC link; and an inverterthat converts a DC voltage between the first DC power supply line andthe second DC power supply line to an output AC voltage, the converterapplying a rectified voltage obtained by full-wave rectifying thesingle-phase AC voltage to the DC link such that the first DC powersupply line is higher in potential than the second DC power supply line,the control device comprising: a duty generating unit that generates arectification duty which is a duty at which a first current flows fromthe rectifying converter to the DC link and a discharge duty which is aduty at which a second current flows from the power buffer circuit tothe DC link; and an inverter controller that outputs an inverter controlsignal that controls an operation of the inverter on the basis of therectification duty, the discharge duty, and a command value of a voltageoutput from the inverter, the rectification duty being set on the basisof a product of a first ratio of a predetermined voltage to an amplitudeof the single-phase AC voltage and an absolute value of a sinusoidalvalue of a phase of the single-phase AC voltage, and a second ratio of athird current to a DC current flowing through the inverter, the thirdcurrent being a fourth current input to the power buffer circuit fromthe DC link or an amount of reduction of a harmonic component of thefourth current.
 2. The control device for the direct power converteraccording to claim 1, wherein the rectifying converter applies therectified voltage obtained by full-wave rectifying the single-phase ACvoltage to the DC link such that the first DC power supply line ishigher in potential than the second DC power supply line, the thirdcurrent is the fourth current, the power buffer circuit includes adischarge circuit including a capacitor and a first switch that isconnected in series to the capacitor between the first DC power supplyline and the second DC power supply line and is closer to the first DCpower supply line than the capacitor is, and a charge circuit thatcharges the capacitor, the control device outputs a discharge switchsignal that brings the first switch into conduction on the basis of thedischarge duty, and the discharge duty is set on the basis of a productof a third ratio of the predetermined voltage to a voltage across thecapacitor and a cosine value of a value twice the phase of thesingle-phase AC voltage, and a product of a fourth ratio of theamplitude to the voltage across the capacitor, the second ratio, and theabsolute value of the sinusoidal value.
 3. The control device for thedirect power converter according to claim 2, wherein the duty generatingunit includes a duty calculating unit that obtains an originalrectification duty, which is a product of the first ratio and theabsolute value of the sinusoidal value, and an original discharge duty,which is a product of the third ratio and a square of a cosine value ofthe phase of the single-phase AC voltage, and a duty correcting unitthat performs correction based on the second ratio to obtain therectification duty from the original rectification duty and thedischarge duty from the original discharge duty.
 4. The control devicefor the direct power converter according to claim 3, wherein the chargecircuit includes a diode including a cathode connected to the capacitorand an anode, a reactor connected between the first DC power supply lineand the anode, and a second switch connected between the second DC powersupply line and the anode, the control device further comprising aswitch control signal generating unit that generates a control signalthat causes the second switch to be turned on once and turned off oncein one period of the rectified voltage.
 5. The control device for thedirect power converter according to claim 2, wherein the charge circuitincludes a diode including a cathode connected to the capacitor and ananode, a reactor connected between the first DC power supply line andthe anode, and a second switch connected between the second DC powersupply line and the anode, the control device further comprising aswitch control signal generating unit that generates a control signalthat causes the second switch to be turned on once and turned off oncein one period of the rectified voltage.
 6. The control device for thedirect power converter according to claim 1, wherein the rectifyingconverter applies the rectified voltage obtained by full-wave rectifyingthe single-phase AC voltage to the DC link such that the first DC powersupply line is higher in potential than the second DC power supply line,the third current is the amount of reduction of the harmonic componentof the fourth current, the power buffer circuit includes a dischargecircuit including a capacitor and a first switch that is connected inseries to the capacitor between the first DC power supply line and thesecond DC power supply line and is closer to the first DC power supplyline than the capacitor is, and a charge circuit that charges thecapacitor, the control device outputs a discharge switch signal thatbrings the first switch into conduction on the basis of the dischargeduty, and the discharge duty is set on the basis of a product of a thirdratio of the predetermined voltage to a voltage across the capacitor anda square of a cosine value of the phase of the single-phase AC voltage,and a product of a fourth ratio of the amplitude to the voltage acrossthe capacitor, the second ratio, and the absolute value of thesinusoidal value.
 7. The control device for the direct power converteraccording to claim 6, the control device further comprising anamount-of-correction generating unit that obtains the amount ofreduction on the basis of a harmonic component of an input current inputto the rectifying converter.
 8. The control device for the direct powerconverter according to claim 7, wherein the duty generating unitincludes a duty calculating unit that obtains an original rectificationduty, which is a product of the first ratio and the absolute value ofthe sinusoidal value, and an original discharge duty, which is a productof the third ratio and a square of a cosine value of the phase of thesingle-phase AC voltage, and a duty correcting unit that performscorrection based on the second ratio to obtain the rectification dutyfrom the original rectification duty and the discharge duty from theoriginal discharge duty.
 9. The control device for the direct powerconverter according to claim 8, wherein the charge circuit includes adiode including a cathode connected to the capacitor and an anode, areactor connected between the first DC power supply line and the anode,and a second switch connected between the second DC power supply lineand the anode, the control device further comprising a switch controlsignal generating unit that generates a control signal that causes thesecond switch to be turned on once and turned off once in one period ofthe rectified voltage.
 10. The control device for the direct powerconverter according to claim 7, wherein the charge circuit includes adiode including a cathode connected to the capacitor and an anode, areactor connected between the first DC power supply line and the anode,and a second switch connected between the second DC power supply lineand the anode, the control device further comprising a switch controlsignal generating unit that generates a control signal that causes thesecond switch to be turned on once and turned off once in one period ofthe rectified voltage.
 11. The control device for the direct powerconverter according to claim 6, the control device further comprising anamount-of-correction generating unit that obtains the amount ofreduction on the basis of the harmonic component of the fourth current.12. The control device for the direct power converter according to claim11, wherein the duty generating unit includes a duty calculating unitthat obtains an original rectification duty, which is a product of thefirst ratio and the absolute value of the sinusoidal value, and anoriginal discharge duty, which is a product of the third ratio and asquare of a cosine value of the phase of the single-phase AC voltage,and a duty correcting unit that performs correction based on the secondratio to obtain the rectification duty from the original rectificationduty and the discharge duty from the original discharge duty.
 13. Thecontrol device for the direct power converter according to claim 12,wherein the charge circuit includes a diode including a cathodeconnected to the capacitor and an anode, a reactor connected between thefirst DC power supply line and the anode, and a second switch connectedbetween the second DC power supply line and the anode, the controldevice further comprising a switch control signal generating unit thatgenerates a control signal that causes the second switch to be turned ononce and turned off once in one period of the rectified voltage.
 14. Thecontrol device for the direct power converter according to claim 11,wherein the charge circuit includes a diode including a cathodeconnected to the capacitor and an anode, a reactor connected between thefirst DC power supply line and the anode, and a second switch connectedbetween the second DC power supply line and the anode, the controldevice further comprising a switch control signal generating unit thatgenerates a control signal that causes the second switch to be turned ononce and turned off once in one period of the rectified voltage.
 15. Thecontrol device for the direct power converter according to claim 6,wherein the duty generating unit includes a duty calculating unit thatobtains an original rectification duty, which is a product of the firstratio and the absolute value of the sinusoidal value, and an originaldischarge duty, which is a product of the third ratio and a square of acosine value of the phase of the single-phase AC voltage, and a dutycorrecting unit that performs correction based on the second ratio toobtain the rectification duty from the original rectification duty andthe discharge duty from the original discharge duty.
 16. The controldevice for the direct power converter according to claim 15, wherein thecharge circuit includes a diode including a cathode connected to thecapacitor and an anode, a reactor connected between the first DC powersupply line and the anode, and a second switch connected between thesecond DC power supply line and the anode, the control device furthercomprising a switch control signal generating unit that generates acontrol signal that causes the second switch to be turned on once andturned off once in one period of the rectified voltage.
 17. The controldevice for the direct power converter according to claim 6, wherein thecharge circuit includes a diode including a cathode connected to thecapacitor and an anode, a reactor connected between the first DC powersupply line and the anode, and a second switch connected between thesecond DC power supply line and the anode, the control device furthercomprising a switch control signal generating unit that generates acontrol signal that causes the second switch to be turned on once andturned off once in one period of the rectified voltage.